Method and system for frequency offset modulation range division MIMO automotive radar

ABSTRACT

A radar system, apparatus, architecture, and method are provided for generating a transmit reference or chirp signal to produce a plurality of transmit signals having different frequency offsets from the transmit reference signal for encoding and transmission as N radio frequency encoded transmit signals which are reflected from a target and received at a receive antenna as a target return signal that is down-converted to an intermediate frequency signal and converted by a high-speed analog-to-digital converter to a digital signal that is processed by a radar control processing unit which performs fast time processing steps to generate a range spectrum comprising N segments which correspond, respectively, to the N radio frequency encoded transmit signals transmitted over the N transmit antennas.

CROSS-REFERENCE TO RELATED APPLICATIONS

U.S. patent application Ser. No. 16/707,235, entitled “Method and Systemfor Frequency Offset Modulation Range Division MIMO Automotive RadarUsing I-Channel Only Modulation Mixer,” by inventors Ryan Haoyun Wu,Douglas Alan Garrity, and Maik Brett, filed on Dec. 9, 2019, describesexemplary methods and systems and is incorporated by reference in itsentirety.

BACKGROUND OF THE INVENTION Field of the Invention

The present invention is directed in general to radar systems andassociated methods of operation. In one aspect, the present inventionrelates to an automotive radar system formed with multiple-input,multiple-output (MIMO) mono-static (co-located) and multi-static(distributed) radar arrays.

Description of the Related Art

Radar systems may be used to detect the range, velocity, and angle ofnearby targets. With advances in technology, radar systems may now beapplied in many different applications, such as automotive radar safetysystems, but not every radar system is suitable for every application.For example, 77 GHz Frequency Modulation Continuous Wave (FMCW) FastChirp Modulation (FCM) radars are used as with very large MIMO arrays assensors in Advanced Driver Assistance System (ADAS) and autonomousdriving (AD) systems. Since the number of virtual antennas constructedwith the MIMO approach (which equals the product of the number ofphysical transmit and receiver antenna elements) is larger than thetotal number of physical elements, the resulting MIMO array can form alarger aperture, resulting in improved angular resolution. However, MIMOsystems can have difficulty distinguishing between Linear FrequencyModulation (LFM) waveforms transmitted by different transmit antennas.

Existing radar systems have attempted to address these challenges byusing time-division (TD) multiplexing techniques to separate LFMwaveforms from different transmitters in time, thereby separatingsignals originated from distinct transmitters at each receiving channelfor constructing a virtual MIMO array. In particular, existing TD MIMOimplementations are configured to schedule a sequence of transmit chirps(LFM waveforms) by individual transmit antennas one element or subarrayat a time, meaning that the amount of time required to transmit allchirps is increased as the number of transmit antennas is increased.Unfortunately, because the coherent dwell time (i.e., the time durationan echo signal of a target can be coherently integrated on a movingtarget) is usually limited, the number of transmitters that can be usedwith TD-MIMO systems is limited. Another drawback with convention alTD-MIMO systems is that longer frame or chirp sequence durations maylead to multiple-times decrease in the maximum Doppler shift (oreffectively, radial velocity of a target) that can be measured withoutambiguity, again limiting the number of transmitters that may be usedfor TD-MIMO systems. As a result, TD-MIMO systems are typically confinedto using a small number of transmitters (e.g., 3) to construct arelatively small MIMO virtual array. As seen from the foregoing, theexisting radar system solutions are extremely difficult at a practicallevel by virtue of the challenges with achieving the performancebenefits of larger size radars within the performance, design,complexity and cost constraints of existing radar system applications.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention may be understood, and its numerous objects,features and advantages obtained, when the following detaileddescription of a preferred embodiment is considered in conjunction withthe following drawings.

FIG. 1 is a simplified schematic block diagram of a conventional LFMTD-MIMO automotive radar system.

FIG. 2 is a timing diagram illustrating a chirp transmission schedulefor an LFM TD-MIMO automotive radar system.

FIG. 3A is a simplified schematic block diagram of a frequency offsetmodulation LFM range division MIMO automotive radar system implementedwith FOM modulation mixers in accordance with selected first embodimentsof the present disclosure.

FIG. 3B is a simplified schematic block diagram of a frequency offsetmodulation LFM range division MIMO automotive radar system implementedwith FOM modulation mixers in accordance with selected secondembodiments of the present disclosure.

FIGS. 4A-B are simplified diagrams of the design and operation of afrequency offset modulation mixer in accordance with selectedembodiments of the present disclosure.

FIG. 5 is a simplified schematic block diagram of frequency offsetmodulation generator in accordance with a first selected embodiment ofthe present disclosure.

FIG. 6 is simplified schematic block diagram of frequency offsetmodulation generator in accordance with a second selected embodiment ofthe present disclosure.

FIG. 7 depicts a fast-time range FFT spectrum of an I-sample onlyreceiver channel of a conventional LFM automotive radar.

FIG. 8 depicts a fast-time range FFT spectrum of a receiver channel ofan I-sample only for an 8-transmitter frequency offset modulated LFMrange-division MIMO automotive radar in accordance with selectedembodiments of the present disclosure.

FIG. 9 depicts a fast-time range FFT spectrum of a receiver channel ofan I-sample only for a 16-transmitter frequency offset modulated LFMrange-division MIMO automotive radar in accordance with selectedembodiments of the present disclosure.

FIG. 10 depicts a fast-time range FFT spectrum of a receiver channel ofan I/Q sample for a 16-transmitter frequency offset modulated LFMrange-division MIMO automotive radar in accordance with selectedembodiments of the present disclosure.

FIG. 11 is a simplified diagram of an I-branch only frequency offsetmodulation mixer in accordance with selected embodiments of the presentdisclosure.

FIG. 12A depicts a fast-time range FFT spectrum of a frequency offsetmodulation LFM range division MIMO automotive radar system such as shownin FIG. 3A which uses I-branch only FOM mixers with insufficientfrequency offset in combination with I-channel only analog-to-digitalconverter in the receiver.

FIG. 12B depicts a fast-time range FFT spectrum of a frequency offsetmodulation LFM range division MIMO automotive radar system such as shownin FIG. 3B which uses I-branch only FOM mixers with insufficientfrequency offset in combination with I-channel only analog-to-digitalconverter in the receiver.

FIG. 13A depicts a fast-time range FFT spectrum of a frequency offsetmodulation LFM range division MIMO automotive radar system such as shownin FIG. 3A which uses I-branch only FOM mixer range spectrum withsufficient frequency offset in combination with I-channel onlyanalog-to-digital converter in the receiver in accordance with selectedembodiments of the present disclosure.

FIG. 13B depicts a fast-time range FFT spectrum of a frequency offsetmodulation LFM range division MIMO automotive radar system such as shownin FIG. 3B which uses I-branch only FOM mixer range spectrum withsufficient frequency offset in combination with I-channel onlyanalog-to-digital converter in the receiver in accordance with selectedembodiments of the present disclosure with the exception that additionaloffset is required to prevent cancellation of a first transmitter'ssignal.

FIG. 13C depicts a fast-time range FFT spectrum of a frequency offsetmodulation LFM range division MIMO automotive radar system such as shownin FIG. 3B which uses I-branch only FOM mixer range spectrum with afirst frequency offset for the first transmitter and a second, doubledfrequency offset for the remaining transmitters in combination withI-channel only analog-to-digital converter in the receiver in accordancewith selected embodiments of the present disclosure.

FIG. 14A diagrammatically depicts a coherent integration of the sum anddelta components of the range spectrums of a frequency offset modulationLFM range division MIMO automotive radar system such as shown in FIG. 3Awhich uses I-branch only FOM mixers in combination with I-channel onlyanalog-to-digital converter in the receiver in accordance with selectedembodiments of the present disclosure.

FIG. 14B diagrammatically depicts a coherent integration of the sum anddelta components of the range spectrums of a frequency offset modulationLFM range division MIMO automotive radar system such as shown in FIG. 3Bwhich uses I-branch only FOM mixers in combination with I-channel onlyanalog-to-digital converter in the receiver in accordance with selectedembodiments of the present disclosure.

FIG. 15 diagrammatically depicts a coherent integration of the sum anddelta components of the range spectrums of a frequency offset modulationLFM range division MIMO automotive radar system such as shown in FIG. 3Bwhich uses I-branch only FOM mixers in combination with I-channel onlyanalog-to-digital converter in the receiver in accordance with selectedembodiments of the present disclosure.

FIG. 16A depicts a fast-time range FFT spectrum of a frequency offsetmodulation LFM range division MIMO automotive radar system such as shownin FIG. 3A which uses I-branch only FOM mixer range spectrum withsufficient frequency offset in combination with PQ-channelanalog-to-digital converter in the receiver in accordance with selectedembodiments of the present disclosure.

FIG. 16B depicts a fast-time range FFT spectrum of a frequency offsetmodulation LFM range division MIMO automotive radar system such as shownin FIG. 3B which uses I-branch only FOM mixer range spectrum withinsufficient frequency offset for the first transmitter in combinationwith PQ-channel analog-to-digital converter in the receiver inaccordance with selected embodiments of the present disclosure.

FIG. 16C depicts a fast-time range FFT spectrum of a frequency offsetmodulation LFM range division MIMO automotive radar system such as shownin FIG. 3B which uses I-branch only FOM mixer range spectrum withsufficient frequency offset for the first transmitter in combinationwith PQ-channel analog-to-digital converter in the receiver inaccordance with selected embodiments of the present disclosure.

FIG. 17 is a simplified schematic block diagram of a frequency offsetmodulation LFM range division MIMO automotive radar system implementedwith fast-time phase shifters in accordance with selected embodiments ofthe present disclosure.

FIG. 18 depicts a fast-time range FFT spectrum for two time slots of areceiver channel of an PQ sample for an 8-transmitter MIMO automotiveradar which employs both time-division and frequency offset modulatedLFM range-division radar techniques in accordance with selectedembodiments of the present disclosure.

FIG. 19 illustrates a simplified flow chart showing the logic for usingfrequency offset modulation techniques to form a virtually large MIMOradar arrays.

DETAILED DESCRIPTION

A frequency offset modulation range and time division MIMO radar system,hardware circuit, system, architecture, and methodology are describedfor combining linear frequency modulation (LFM) time-division (TD) MIMOwith frequency offset modulation (FOM) range division MIMO to constructvery large MIMO arrays for use with frequency modulation continuous wave(FMCW) radars. In selected embodiments, a signal processing methodologyand apparatus are disclosed for mixing the LFM waveform (transmit chirp)at each transmit channel with different frequency offset signals (e.g.,Δf, 2Δf, etc.) using a frequency offset mixer with increased ADCsampling rate to allow the separation of transmitters' signals onreceive in the range spectrum, thereby enabling very large MIMO arrayformation at the receiver. With each transmit channel transmitting adifferent frequency offset modulation LFM signal, the receiver canprocess and separate the transmit channel signals in the fast-timeFourier or the range domain, thereby defining an LFM range-division (RD)MIMO approach for differentiating between transmit channel signals. Inselected embodiments, the frequency offset mixer may be implemented withan I/Q channel modulation mixer, an I-channel only modulation mixer, ora Q-channel only modulation mixer to implement a spectrum-coherentintegration approach. In embodiments where the FOM mixer is implementedwith an I-channel only modulation mixer or a Q-channel only modulationmixer with a spectrum domain coherent integration approach, thecomplexity of the hardware implementations is greatly reduced. In otherembodiments, a signal processing methodology and apparatus are disclosedfor frequency shifting the LFM waveform (transmit chirp) at eachtransmit channel with different fast-time phase shifters prior totransmit filtering and amplification, thereby enabling very large MIMOarray formation at the receiver. With each transmit channel transmittinga different frequency offset modulation LFM signal that is generatedwith a fast-time phase shifter, there is no requirement of a frequencyoffset mixer at each transmit channel, and a high speedanalog-to-digital converter at the receiver can process and separate thetransmit channel signals in the fast-time Fourier or the range domain,thereby defining an LFM range-division (RD) MIMO approach fordifferentiating between transmit channel signals. Because of thesimultaneous transmissions of FOM RD MIMO, the frame duration can bekept short, thereby avoiding the problem associated with prolonged frameduration of TD MIMO systems. The resulting FOM RD MIMO virtual array ismuch larger than the conventional TD-MIMO approach, and can provide highangular resolution performance. In selected embodiments, a signalprocessing methodology and apparatus are disclosed for combining the FOMRD MIMO approach with TD MIMO approach, such as by defining multipletransmit time slots such that alternating transmit channels (e.g., evennumber transmitter) are active in a first time slot and are suppressedin a second time slot. In this configuration, for each transmitter, itsadjacent range spectrum segment is vacant, thereby enabling strongbeyond-the-range targets to be correctly detected without imposingtarget interference. By providing hardware and software solutions forusing frequency offset modulation in combination with time-divisiontechniques to transmit LFM waveforms from multiple transmit channels,the disclosed frequency offset modulation range and time division MIMOradar system and methodology efficiently provide a MIMO virtual arrayhaving an aperture that is many times larger than the total physicalapertures combined, thereby achieving better sensitivity, finer angularresolution, and low false detection rate.

In the context of the present disclosure, it will be appreciated thatradar systems may be used as sensors in a variety of differentapplications, including but not limited to automotive radar sensors forroad safety systems, such as advanced driver-assistance systems (ADAS)and autonomous driving (AD) systems. In such applications, the radarsystems are used to measure the radial distance to a reflecting object,its relative radial velocity, and angle information, and arecharacterized by performance criteria, such as the angular resolution(the minimum distance between two equal large targets at the same rangeand range rate (or radial velocity) resolution cell which a radar isable to distinguish and separate to each other), sensitivity, falsedetection rate, and the like. Typically, frequency modulated continuouswave (FMCW) modulation radars are used to identify the distance,velocity, and/or angle of a radar target, such as a car or pedestrian,by transmitting Linear Frequency Modulation (LFM) waveforms frommultiple transmit antennas so that reflected signals from the radartarget are received at multiple receive antennas and processed todetermine the radial distance, relative radial velocity, and angle (ordirection) for the radar target. However, with current automotivedesigns, a vehicle can include multiple radar transmitters which canoperate independently from one another. As a result, the LFM waveformtransceivers may be configured to implement time-division (TD) MIMOoperations to temporally separate signals originated from distincttransmitters so that a receiving channel can distinctly detect eachsignal and thereby construct a virtual MIMO array.

To illustrate the design and operation of a conventional TD MIMO radarsystem, reference is now made to FIG. 1 which depicts a simplifiedschematic block diagram of a conventional LFM TD-MIMO automotive radarsystem 100 which includes an LFM TD-MIMO radar device 10 connected to aradar controller processor 20. In selected embodiments, the LFM TD-MIMOradar device 10 may be embodied as a line-replaceable unit (LRU) ormodular component that is designed to be replaced quickly at anoperating location. Similarly, the radar controller processor 20 may beembodied as a line-replaceable unit (LRU) or modular component. Althougha single or mono-static LFM TD-MIMO radar device 10 is shown, it will beappreciated that additional distributed radar devices may be used toform a distributed or multi-static radar. In addition, the depictedradar system 100 may be implemented in integrated circuit form with theLFM TD-MIMO radar device 10 and the radar controller processor 20 formedwith separate integrated circuits (chips) or with a single chip,depending on the application.

Each radar device 10 includes one or more transmitting antenna elementsTX_(i) and receiving antenna elements RX_(j) connected, respectively, toone or more radio-frequency (RF) transmitter (TX) units 11 and receiver(RX) units 12. For example, each radar device (e.g., 10) is shown asincluding individual antenna elements (e.g., TX_(1,i), RX_(1,j))connected, respectively, to three transmitter modules (e.g., 11) andfour receiver modules (e.g., 12), but these numbers are not limiting andother numbers are also possible, such as four transmitter modules 11 andsix receiver modules 12, or a single transmitter module 11 and/or asingle receiver modules 12. Each radar device 10 also includes a chirpgenerator 112 which is configured and connected to supply a chirp inputsignal to the transmitter modules 11. To this end, the chirp generator112 is connected to receive a separate and independent local oscillator(LO) signal 110 and a chirp start trigger signal 111, though delays arelikely to be different due to the signal path differences andprogrammable digital delay elements in the signal paths. Chirp signals113 are generated and transmitted to multiple transmitters 11, usuallyfollowing a pre-defined transmission schedule, where they are filteredat the RF conditioning module 114 and amplified at the power amplifier115 before being fed to the corresponding transmit antenna TX_(1,i) andradiated. By sequentially using each transmit antenna TX_(1,i) totransmit successive pulses in the chirp signal 113, each transmitterelement 11 operates in a time-multiplexed fashion in relation to othertransmitter elements because they are programmed to transmit identicalwaveforms on a temporally separated schedule.

The radar signal transmitted by the transmitter antenna unit TX_(1,i),TX_(2,i) may by reflected by an object, and part of the reflected radarsignal reaches the receiver antenna units RX_(1,i) at the radar device10. At each receiver module 12, the received (radio frequency) antennasignal is amplified by a low noise amplifier (LNA) 120 and then fed to amixer 121 where it is mixed with the transmitted chirp signal generatedby the RF conditioning unit 113. The resulting intermediate frequencysignal is fed to a first high-pass filter (HPF) 122. The resultingfiltered signal is fed to a first variable gain amplifier 123 whichamplifies the signal before feeding it to a first low pass filter (LPF)124. This re-filtered signal is fed to an analog/digital converter (ADC)125 and is output by each receiver module 12 as a digital signal D1. Thereceiver module compresses target echo of various delays into multiplesinusoidal tones whose frequencies correspond to the round-trip delay ofthe echo.

The radar system 100 also includes a radar controller processing unit 20that is connected to supply input control signals to the radar device 10and to receive therefrom digital output signals generated by thereceiver modules 12. In selected embodiments, the radar controllerprocessing unit 20 may be embodied as a micro-controller unit (MCU) orother processing unit that is configured and arranged for signalprocessing tasks such as, but not limited to, target identification,computation of target distance, target velocity, and target direction,and generating control signals. The radar controller processing unit 20may, for example, be configured to generate calibration signals, receivedata signals, receive sensor signals, generate frequency spectrumshaping signals (such as ramp generation in the case of FMCW radar)and/or register programming or state machine signals for RF (radiofrequency) circuit enablement sequences. In addition, the radarcontroller processor 20 may be configured to program the modules 11 tooperate in a time-division fashion by sequentially transmitting LFMchirps for coordinated communication between the transmit antennasTX_(1,i), RX_(1,j). The result of the digital processing at the radarcontroller processing unit 20 is that the digital domain signals D1 areprocessed for the subsequent fast-time range FFT 21, slow-time DopplerFFT 22, constant false alarm rate (CFAR) target detection 23, spatialangle estimation 24, and target tracking processes 25, with the resultbeing output 26 to other automotive computing or user interfacingdevices for further process or display.

To illustrate an example of time division transmission of radar transmitsignals, reference is now made to FIG. 2 which depicts a timing diagramillustration 200 of a chirp transmission schedule for an LFM TD-MIMOautomotive radar system. As depicted, each transmitter (e.g., TX₁, etc.)is programed to take turns transmitting one chirp (e.g., 201) of asequence of chirps 201-206. This temporal separation of chirptransmission by each transmit antenna allows the separation oftransmitters at the receiving end by simply associating the receivedsignal with the scheduled transmitter. The ability to separatetransmitters in the received signal is a prerequisite of the MIMO radarapproach, which is routinely used in automotive radars for constructinga virtually large antenna array aperture compared to the physicalaperture of the transmit and receive antennas. The larger apertureconstructed virtually via MIMO provides better angular resolutionperformance which is required by many advanced driver assistance system(ADAS) and autonomous driving (AD) applications. The use of atransmitter schedule to divide the time-domain resources amongst thetransmitters when forming a virtual MIMO array is referred to as a timedivision (TD) MIMO approach.

Since the TD-MIMO approach provides a relatively straightforward way toseparate transmitters with little or no leakage, it is routinely used inautomotive radar applications. However, the requirement of dividing timebetween resources means that a much longer frame duration is required tocomplete the transmission of all chirps for each transmitter. If theprolonged frame duration is longer than the duration a target stayswithin a single range resolution cell, any range migration by the targetcan degrade the subsequent digital Doppler coherent integrationprocessing and angle estimation, thereby adversely impacting measurementperformance.

Another drawback with conventional TD-MIMO approaches is the increase inthe duration of the pulse repetition intervals (PRI) between adjacentpulses of the same transmitter. In particular, with each transmitter(e.g., TX₁-TX_(N)) being scheduled to take its turn to transmit theirfirst pulses (e.g., 201-202) before beginning the sequentialtransmission of the second pulses (e.g., 203-204), and so on until thelast pulses (e.g., 205-206) are transmitted, the pulse repetitioninterval (PRI) 202 between two adjacent pulses of the same transmitteris also prolonged. Because the maximum unambiguous Doppler shiftmeasurable by the chirp sequence is inversely related to the PRI, alengthened PRI results in reduced maximum unambiguous Dopplerperformance. As a result, the maximum number of transmitters that can beused for TD-MIMO operation is limited. For typical road use, up to 3transmitters may be used for TD MIMO without unacceptable performancedegradation.

To address these limitations from conventional solutions and othersknown to those skilled in the art, reference is now made to FIG. 3Awhich depicts a simplified schematic block diagram of a frequency offsetmodulation LFM range division MIMO automotive radar system 300A whichincludes an LFM RD-MIMO radar device 330A having a transmit module 310Aand receiver module 320 which are connected and configured to transmitand receive LFM waveforms 302A, 303A for reflection by a target 301under control of a radar controller processor (not shown). In selectedembodiments, the LFM RD-MIMO radar device 330A and/or radar controllerprocessor may be embodied as a line-replaceable unit (LRU) or modularcomponent that is designed to be replaced quickly at an operatinglocation. In addition and as described hereinbelow, the LFM RD-MIMOradar device 330A may also be configured to perform time-divisionmultiplexing of the transmitted LFM waveforms 302A, 303A to implement acombined time-division and range-division MIMO scheme to separate thetransmitters not only in the range domain, but also in the time domain.

Each radar device 330A includes one or more transmitting antennaelements TX_(i) and at least a first receiving antenna element RXconnected, respectively, to one or more radio-frequency (RF) transmitmodules 310A and receive module 320. At each transmit module 310A, atransmit channel circuit is provided for each transmit antenna. Forexample, a first transmit channel circuit includes a first RFconditioning module 311 and power amplifier 312 connected to a firsttransmit antenna TX₁, a second transmit channel circuit includes asecond RF conditioning module 314 and power amplifier 315 connected to asecond transmit antenna TX₂, and so on with the Nth transmit channelcircuit including an Nth RF conditioning module 317 and power amplifier318 connected to the Nth transmit antenna TX_(N).

In addition, each radar device 330A includes a chirp generator 304 whichis configured and connected to supply a chirp input signal 305 to thedifferent transmit channel circuits 311/312, 314/315, 317/318 in thetransmitter module(s) 310A. However, instead of providing the chirpinput signal 305 directly to all of the transmit channel circuits, theradar device 330A also includes a frequency offset generator 306A andfrequency offset modulator (FOM) mixers 313, 316 which are connected toshift or offset the frequency of each transmitted LFM waveform 302A by adifferent integer multiple of a frequency offset (Δf). To this end, thefirst transmit channel circuit 311/312 may be connected to directlyreceive the chirp input signal 305. However, the second transmit channelcircuit 314/315 may include an FOM mixer 313 which is connected as anI/Q mixer to apply a first frequency offset signal Δf to the chirp inputsignal 305 before being filtered and amplified by the second transmitchannel circuit 314/315 for transmission over the antenna TX₂. Insimilar fashion, the remaining transmit channel circuits (e.g., 317/318)may include an FOM mixer (e.g., 316) which is connected as an I/Q mixerto apply a unique frequency offset signal (e.g., (N−1)Δf) to the chirpinput signal 305 before being filtered, amplified, and radiated by theantenna (e.g., TX_(N)).

With the frequency offset generator 306A connected to receive thereference local oscillator signal LO, a plurality of predefinedfrequency offset signals 307A may be generated, such as by generating adifferent integer multiple of a frequency offset (Δf) for eachtransmitter channel circuit which is connected to receive a frequencyoffset. By shifting the center frequency of LFM waveform at eachtransmitter TX_(i) according to a predefined offset frequency unique toeach transmitter, frame durations can be reduced (as compared to TDschemes) to prevent range migration effect from degrading the Dopplerand angle processing. In selected embodiments, the amount of frequencyoffsets should be sufficient such that no targets within the maximumdetection range of the radar from any two transmitters overlaps in therange domain. Thus, the amount of range shift corresponds to the amountof frequency offset imposed on each transmitter and is at least theamount of the maximum detection range of the radar.

For each transmit channel circuit except the first transmit channelcircuit 311/312, the frequency offset generator 306A generates afrequency offset tone based on a fundamental offset frequency Δf that isderived from the LO signal and multiplied by an integer (e.g., 1, 2, 3,4, . . . N−1), with a different integer value being used for eachtransmit channel circuit. For example, the frequency offset generator306A generates a first frequency offset tone 1×Δf that is supplied tothe FOM mixer 313 for the second transmit channel circuit 314/315.Likewise, the frequency offset generator 306A generates a secondfrequency offset tone 2×Δf that is supplied to the FOM mixer for thethird transmit channel circuit, and so on, with the frequency offsetgenerator 306A generating a last frequency offset tone (N−1)×Δf that issupplied to the FOM mixer 316 for the last transmit channel circuit317/318. In this way, the frequency offset generator 306A provides N−1offset tones for the transmit channel circuits. If desired, anadditional 2NΔf Hz component can be generated to drive the ADC if thesampling rate if f_(s)=2NΔf.

As will be appreciated, a variety of different configurations may beused to deploy the frequency offset generator and FOM mixers forpurposes of frequency-shifting each transmit channel. For example,reference is now made to FIG. 3B which depicts a simplified schematicblock diagram of a frequency offset modulation LFM range division MIMOautomotive radar system 300B which includes an LFM RD-MIMO radar device330B having a transmit module 310B and receive module 320 which areconnected and configured to transmit and receive LFM waveforms 302B,303B for reflection by a target 301 under control of a radar controllerprocessor (not shown). As depicted, the LFM RD-MIMO radar device 330B issimilar to the LFM RD-MIMO radar device 330A shown in FIG. 3B, exceptthat the chirp input signal 305 is not directly provided to any of thetransmit channel circuits. Instead, the frequency offset generator 306Band frequency offset modulator (FOM) mixers 313, 316 are connected toshift or offset the frequency of the chirp input signal 305 by adifferent integer multiple of a frequency offset (Δf) before beingprovided to any transmit channel circuit. In particular, the firsttransmit channel circuit 314/315 includes an FOM mixer 313 which isconnected as an I/Q mixer to apply a first frequency offset signal Δf tothe chirp input signal 305 before being filtered and amplified by thefirst transmit channel circuit 314/315 for transmission over the antennaTX₁. In similar fashion, the remaining transmit channel circuits (e.g.,317/318) include an FOM mixer (e.g., 316) which is connected as an I/Qmixer to apply a unique frequency offset signal (e.g., (N)Δf) to thechirp input signal 305 before being filtered and amplified by theantenna (e.g., TX_(N)). With this configuration, the frequency offsetgenerator 306B is connected to receive the reference local oscillatorsignal LO and to generate therefrom a plurality of predefined frequencyoffset signals 307B as a different integer multiple of a frequencyoffset (Δf) for each corresponding transmitter channel circuit. Byshifting the center frequency of LFM waveform 302B at each transmitterTX_(i) according to a predefined offset frequency unique to eachtransmitter, the combined reflected LFM waveforms 303B are received andprocessed by the receiver module 320 in the substantially the same wayas described hereinbelow.

To ensure sufficient range space is available for division amongst alltransmitters, a much longer system-describable unambiguous range extentmust be provided by the fast-time sampling. Because the maximumunambiguous range extent is inversely related to the fast-time samplinginterval, a faster analog-to-digital converter (ADC) is employed at eachreceive channel. For example, a conventional TD-MIMO FCM radar may use a40 mega-samples-per-second (Msps) ADC, but an LFM MIMO radar deviceusing frequency offset modulation to enable N-transmitter MIMO operationshould use an ADC sampling rate that is increased to N×40 Msps. Notealso that depending on the radar system requirements and also upon theactual performance of the low-pass filter that directly precedes theADC, the sample rate of the ADC may need to be increased beyond N×40Msps. As a result, the fast-time FFT processing can divide the spectruminto N consecutive segments, with each being associated with acorresponding transmitter. Because the transmitters are separated ordivided in the range domain and the waveform is based on LFM, theapproach can also be referred to as the LFM range-division (RD) MIMOapproach.

With the frequency offset signals 307A, 307B applied to the chirp inputsignal 305 by a bank of FOM mixers 313, 316 at the transmitter channelcircuits before transmission on the transmit antennas TX₁-TX_(N), thecombined reflected LFM waveforms 303A, 303B are received and processedby the receiver module 320. In particular, the receive antenna RXreceives the combined reflected LFM waveforms 303A, 303B which are thenamplified by the low noise amplifier (LNA) 321. At the I/Q mixer 322,the amplified receive signal is mixed with the reference chirp signal305 before being conditioned for digital conversion by the high passfilter 323, variable gain amplifier 324, low pass filter 325, andanalog-to-digital converter 326.

In some cases, the signal radiated from the transmitters TX₁-TX_(N) isnot sufficiently isolated from the receiver which can result inundesirable interference that presents itself as an artificial target atnear-zero range. This is known as transmitter-to-receiver spill-overinterference. In selected embodiments of the present disclosure, ananalog-domain tunable and configurable notch filter bank circuit may besubstituted for the high pass filter 323 for filtering out thezero-range interference in the fast-time spectrum of each transmitter.Because it is not a target return, its presence interferes with thedetection of valid targets that are close in distance due the spectralskirts caused by the phase noise of the system. In addition, reflectivestructures (e.g., the bumper) of the car around the radar result in aclose-in echo (with nearly zero range) which interferes with thedetection of valid targets. Such interference can be suppressed byapplying analogue filters and/or digital filters. In FMCW radar designswhere an analog high-pass filter (HPF) with a tunable pass-bandfrequency is employed after the chirp mixer 322 to suppress thezero-range interference signals, such a single high-pass filter cannotfilter out all of the multiple zero-range signals in the received FOMsignal occurring at multiple non-zero frequencies. To suppress theseinterference signals, a bank of notch filters (also known as a combfilter) is employed where each notch filter is tuned to a correspondingzero-range frequency, thereby providing a FOM LFM RD MIMO radar withspill-over interference cancellation using a notch filter in the receivepath.

To provide an improved understanding of how the FOM mixers 313, 316shift the frequency of the reference chirp, reference is now made toFIG. 4A which is a simplified diagram 400 of the design and operation ofa frequency offset modulation mixer 401 which shifts the frequency ofthe offset tone mixed with the reference chirp signal. Denoting thereference chirp signal as s_(b)(t)sin(2πf₀ t), it will be understoodthat sin(2πf₀ t) represents the carrier with a center frequency of f₀,and s_(b) (t) denotes the baseband chirp waveform that is modulated bythe carrier tone. Since the chirp bandwidth is much smaller than thecarrier frequency, it can be treated as a narrow band signal. To shiftthe reference chirp's carrier frequency by f_(Δ), an offset tone signalsin(2πf_(Δ)t) is generated for input to the FOM mixer 401. As indicatedby the cascaded mixer symbol, the FOM mixer 401 is a I/Q mixer formixing the LFM waveform of the reference chirp signal (s_(b)(t)sin(2πf₀t)) with a fixed frequency of the offset tone (sin(2πf_(Δ)t)).

To provide additional details for an improved understanding of selectedembodiments of the present disclosure, reference is now made to FIG. 4Bwhich is a simplified diagram of the design and operation of a frequencyoffset modulation mixer 451. As depicted, the reference chirp inputsignal (sin(2πf₀ t)) is first split into I and Q component branches,with one branch shifted by 90 degrees (at shifter 454) to generate the Qcomponent (cos(2πf₀ t))) for mixing with the I component of the offsettone (sin(2πf_(Δ)t)) at the I/Q mixer 453 to produce the mixer outputx₂(t). In similar fashion, the I component of the reference chirp inputsignal (sin(2πf₀ t)) is mixed with the 90-degree phase-shifted offsettone signal (cos(2πf_(Δ)t)) which is generated by the 90 degree shifter452 at the I/Q mixer 455 to produce the mixer output x₁(t). Finally, theI/Q mixer circuit 456 combines the outputs x₁(t), x₂(t) of the twomixers 453, 455 to form the final frequency offset modulated chirpsignal (sin(2π(f₀+f_(Δ))t)).

To demonstrate that the sum of the two branches is the frequency-offsetmodulated chirp signal, the following derivation may be established:x ₁(t)=sin(2πf ₀ t)cos(2πf _(Δ) t)=0.5(sin(2π(f ₀ +f _(Δ))t)+sin(2π(f ₀−f _(Δ))t))x ₂(t)=cos(2πf ₀ t)sin(2πf _(Δ) t)=0.5(sin(2π(f ₀ +f _(Δ))t)−sin(2π(f ₀f _(Δ))t))x ₁(t)+x ₂(t)=sin(2π(f+f _(Δ))t)

For each transmit channel circuit, the frequency offset generator 306generates a frequency offset tone based on a fundamental offsetfrequency Δf that is derived from the LO signal and multiplied by aninteger (e.g., 1, 2, 3, 4, . . . N−1), with a different integer valuebeing used for each transmit channel circuit. For example, the frequencyoffset generator 306A generates a first frequency offset tone 1×Δf thatis supplied to the FOM mixer 313 for the second transmit channel circuit314/315. Likewise, the frequency offset generator 306A generates asecond frequency offset tone 2×Δf that is supplied to the FOM mixer forthe third transmit channel circuit, and so on, with the frequency offsetgenerator 306A generating a last frequency offset tone (N−1)×Δf that issupplied to the FOM mixer 316 for the last transmit channel circuit317/318. In this way, the frequency offset generator 306A provides N−1offset tones for the transmit channel circuits. If desired, anadditional 2NΔf Hz component can be generated to drive the ADC if thesampling rate if f_(s)=2NΔf. Note also that depending on the radarsystem requirements and also upon the actual performance of the low-passfilter 325 that directly precedes the ADC, the sample rate of the ADCmay need to be increased beyond f_(s)=2NΔf.

While any suitable frequency offset generator circuit arrangement may beused, reference is now made to FIG. 5 which depicts a simplifiedschematic block diagram of frequency offset modulation generator 500 inaccordance with a first selected embodiment of the present disclosure.As depicted, the frequency offset modulation generator 500 is connectedto receive the fundamental offset frequency tone (sin(2πf_(Δ)t)) as aninput, and to output the received frequency tone as a first outputoffset frequency tone (sin(2πf_(Δ)t)). The frequency offset modulationgenerator 500 also includes a plurality of frequency multiplier circuits502-506, each providing a different integer multiplier function togenerate a corresponding output. For example, a first frequencymultiplier circuit 502 provides a 2× multiplier function to generate asecond output offset frequency tone (sin(2πf_(Δ)t)). In addition, asecond frequency multiplier circuit 503 provides a 3× multiplierfunction to generate a third output offset frequency tone(sin(2π3f_(Δ)t)), a third frequency multiplier circuit 504 provides a 4×multiplier function to generate a third output offset frequency tone(sin(2π4f_(Δ)t)), and so on, with the a (N−1)th frequency multipliercircuit 505 providing an (N−1)x multiplier function to generate the(N−1)th output offset frequency tone (sin((N−1)π2f_(Δ)t)). In selectedembodiments, the frequency offset modulation generator 500 may alsoinclude an additional frequency multiplier 506 that provides a 2Nmultiplier function to the generate the sampling output offset frequencytone (sin(2π2Nf_(Δ)t)) for driving the ADC in the receiver module if thesampling rate if f_(s)=2NΔf. Note also that depending on the radarsystem requirements and also upon the actual performance of the low-passfilter 325 that directly precedes the ADC, the sample rate of the ADCmay need to be increased beyond f_(s)=2NΔf.

While the frequency offset tones can be generated from the fundamentaloffset frequency using a plurality of multipliers, it will beappreciated that other approaches for generating the offset frequenciesmay be used. For example, reference is now made to FIG. 6 which depictsa simplified schematic block diagram of frequency offset modulationgenerator 600 in accordance with a second selected embodiment of thepresent disclosure wherein the FOM tones are generated from the samplingfrequency of the ADC LO at f_(s) Hz. As depicted the frequency offsetmodulation generator 600 is connected to receive the ADC samplingfrequency tone (sin(2πf_(s)t)) as an input to a first frequency dividercircuit 601 which divides the input frequency by a factor of 2N foroutput as a first output offset frequency tone (sin(2πf_(Δ)t)). Thefrequency offset modulation generator 600 also includes a plurality offrequency multiplier circuits 602-605, each providing a differentinteger multiplier function to generate a corresponding output. Forexample, a first frequency multiplier circuit 602 provides a 2×multiplier function to the output from the first frequency dividercircuit 601, thereby generating a second output offset frequency tone(sin(2πf_(Δ)t)). In addition, a second frequency multiplier circuit 603provides a 3× multiplier function to the output from the first frequencydivider circuit 601, thereby generating a third output offset frequencytone (sin(2π3f_(Δ)t)), and a third frequency multiplier circuit 604provides a 4× multiplier function to the output from the first frequencydivider circuit 601, thereby generating a third output offset frequencytone (sin(2π4f_(Δ)t)), and so on, with the a (N−1)th frequencymultiplier circuit 605 providing an (N−1)x multiplier function togenerate the (N−1)th output offset frequency tone (sin((N−1)πf_(Δ)t)).

For a contextual understanding of the design and arrangement of arange-divided transmitter signal spectrum produced by a frequency offsetmodulation LFM range division MIMO automotive radar system, reference isnow made to FIG. 7 which depicts an example fast-time range FFT spectrum700 of an I-sample only receiver channel of a conventional LFMautomotive radar having a designed 2π range of 400 m which correspondsto a 2π IF frequency of 40 MHz. In the example, the potential rangespectrum 700 of such a radar is generated from a single transmit antennaTx1 by sampling only the real part (I samples) of the mixer output atthe receiver for use in producing the spectrum. Given real samples only,it can be seen that the spectrum is conjugate symmetric around the zerofrequency, in which case the usable range extent is between 0 and 200 m,and the maximum unambiguous range of this radar is designed to be 200 m.It will also be appreciated that any simultaneous transmission frommultiple transmit antennas of the same of LFM waveforms will result inoverlapping signals in the same range FFT spectrum segment, making itimpossible to extract individual transmit channel information. Whilethis information extraction problem can be addressed by temporallyseparating the LFM waveform transmissions from each transmit antenna,such a time-division scheme by itself has the performance limitationsand drawbacks noted hereinabove.

To address these limitations and others known to those skilled in theart, there is disclosed herein a frequency offset modulation (FOM)approach for use with LFM automotive radar systems to separateindividual transmit channel information within the fast-time range FFTspectrum at the receiver by having the transmitter mix a uniquefrequency offset signal with the chirp signal of each transmit channel.To illustrate the resulting transmit signal spectrum, reference is nowmade to FIG. 8 which depicts a fast-time range FFT spectrum 800 of areceiver channel of an I-sample only mixer which receives an8-transmitter frequency offset modulated LFM range-division MIMOautomotive radar in accordance with selected embodiments of the presentdisclosure. Based on the waveform configuration shown in FIG. 7 , theapplication of frequency offset modulation to support 8 transmitters forRD-MIMO will require that the sampling rate be increased to at least 320MHz so that each transmitter occupies a portion of the spectrum withoutoverlapping each other. As depicted with the darker lines, thetransmitter channels Tx1-Tx8 occupy consecutive 20 MHz spectrum segmentsbetween 0 and 160 MHz, and as indicated with the gray lines, the rest ofthe spectrum 800 is conjugate symmetrically redundant due to the use ofreal sample only data. In this case, the offset frequency f_(Δ) betweentwo adjacent transmitters is 20 MHz. Again, the use of only real samplesmeans that the spectrum is conjugate symmetric around the zerofrequency, and the usable range extent for each transmitter Tx1-Tx8 isbetween 0 and 200 m, with each transmitter's range extent being shiftedor offset from one another in the range FFT spectrum 800 (e.g., 0m≤Tx1<200 m, 200 m≤Tx2<400 m, 400 m≤Tx3<600 m, etc.). Based on therange-spectrum division arrangement whereby individual transmittersoccupy distinct portions of the spectrum 800, the fast-time samplesassociated with distinct transmitters may be recovered at the receivermodule for subsequent processing to construct the MIMO virtual array.

To provide another example transmitter signal spectrum for an LFM rangedivision MIMO automotive radar system, reference is now made to FIG. 9which depicts a fast-time range FFT spectrum 900 of a receiver channelof an I-sample only for a 16-transmitter frequency offset modulated LFMrange-division MIMO automotive radar in accordance with selectedembodiments of the present disclosure. Assuming the same waveformconfiguration as before, the ADC sampling rate must be increased to atleast 640 MHz so that each transmitter occupies at least 20 MHz of thespectrum without overlapping each other. Note also that depending on theradar system requirements and also upon the actual performance of thelow-pass filter 325 that directly precedes the ADC, the sample rate ofthe ADC may need to be increased beyond f_(s)=2NΔf. As depicted with thedarker lines, the transmitter channels Tx1-Tx16 occupy consecutive 20MHz spectrum segments between 0 and 320 MHz, and as indicated with thegray lines, the rest of the spectrum 900 is conjugate symmetricallyredundant due to the use of real sample only data.

As will be appreciated, other sampling arrangements can be implementedwith FOM LFM RD-MIMO automotive radar systems to provide a fast-timerange division FFT spectrum. For example, both the real and imaginary (Iand Q) samples may be used for producing the spectrum. To provide anexample of transmitter signal spectrum generated from I and Q samples,reference is now made to FIG. 10 which depicts a fast-time range FFTspectrum 1000 of a receiver channel of an I/Q sample for a16-transmitter frequency offset modulated LFM range-division MIMOautomotive radar in accordance with selected embodiments of the presentdisclosure. By virtue of using both I and Q samples, the sampling rateof each of the I and Q channel ADCs can be reduced to 320 MHz whilestill allowing the 16 transmitters to fit within the 320 MHz swath ofspectrum while maintaining at least 20 MHz of the spectrum for eachtransmitter without overlapping each other. In particular, thetransmitter channels Tx1-Tx16 occupy consecutive 20 MHz spectrumsegments between 0 and 320 MHz. However, this arrangement requiresbalanced I/Q mixers and two ADCs (one for I samples and one for Qsamples) which may be costlier than the I-sample only implementation forachieving similar performance.

While selected embodiments of the FOM MIMO scheme are described withreference to using I/Q modulation mixers to combine offset frequencysignals (f_(t), 2f_(t), 3f_(t), etc.) with the reference chirp signal,it will be appreciated that some I/Q modulators impose significanthardware costs and complexity, such as the requirement for implementingphase shifters and/or for avoiding unbalanced I/Q mixing that can arisewhen there is misalignment between the amplitude and phase of the I andQ channels. To provide additional details for an improved understandingof selected single channel modulation mixer embodiments of the presentdisclosure, reference is now made to FIG. 11 which depicts a simplifieddiagram of an I-branch only frequency offset modulation mixer 1100 whichshifts the carrier frequency of the reference chirp signal sin(2πf₀ t)by the offset tone signal sin(2πf_(Δ)t) which are both input to theI-channel only FOM mixer 1101. In this example, the frequency f₀represents the chirp signal's instantaneous frequency, and the frequencyf_(Δ) represents a constant offset frequency. As indicated by the singlemixer symbol, the FOM mixer 1101 is a single channel (e.g., I-channelonly) mixer for mixing the LFM waveform of the reference chirp signalcarrier sin(2πf₀ t)) with a fixed frequency of the offset tone(sin(2πf_(Δ)t)). As will be appreciated, the same operative principlesmay be used to implement a Q-branch only frequency offset modulationmixer (not shown) for shifting the carrier frequency of the referencechirp signal sin(2πf₀ t) by the offset tone signal sin(2πf_(Δ)t).

In the depicted example of an I-channel only FOM mixer 1101, thereference chirp input signal carrier (sin(2πf₀ t)) is provided as anI-channel for mixing with the I component of the offset tone(sin(2πf_(Δ)t)) at the I-channel only mixer 1102 to produce the mixeroutput x₁(t) which includes a first down-shifted delta component (e.g.,0.5(cos(2π(f₀−f_(Δ))t)) and a second up-shifted sum component (e.g.,−0.5(cos(2π(f₀+f_(Δ))t)). To demonstrate that the sum of the twocomponents is the frequency-offset modulated chirp signal, the followingderivation of the mixer output x₁(t) may be established based uponestablished trigonometric identities:

$\begin{matrix}{{x_{1}(t)} = {{0.5{\cos\left( {2{\pi\left( {f_{0} - f_{\Delta}} \right)}t} \right)}} - {{0.5}{\cos\left( {2{\pi\left( {f_{0} + f_{\Delta}} \right)}t} \right)}}}} \\{= {{0.5{\sin\left( {{2{\pi\left( {f_{0} - f_{\Delta}} \right)}t} + {\pi/2}} \right)}} + {{0.5}{{\sin\left( {{2{\pi\left( {f_{0} + f_{\Delta}} \right)}t} - {\pi/2}} \right)}.}}}}\end{matrix}$

With this derivation, it is seen that the I-channel only mixer outputx₁(t) includes a first down-shifted delta component (e.g.,0.5(sin(2π(f₀−f_(Δ))t+π/2)) and a second up-shifted sum component (e.g.,0.5 sin(2π(f₀+f_(Δ))t−π/2)). The sum component shifts the range spectrumof a transmitter by the amount of offset frequency to the positive rangedirection. The delta component shifts the range spectrum of atransmitter by the amount of offset frequency to the negative rangedirection.

As will be appreciated, the mixer output x₁(t) value when the frequencyoffset f_(Δ)=0 will depend on the configuration of the frequency offsetgenerator (e.g., FIG. 3A, 306A or FIG. 3B, 306B) and the FOM mixers inthe transmit modules (e.g., FIG. 3A, 310A or FIG. 3B, 310B). Forexample, with the transmit module configuration shown in FIG. 3A wherethe first transmit channel 311, 312 has no mixer, the mixer outputx₁(t)=sin(2πf₀t), which is out-of-phase with the other channel'sup-shifted sum component by π/2 and therefore requires proper handlingat the receiver. In contrast, with the transmit module configurationshown in FIG. 3B where the each transmit channel includes a mixer, themixer output x₁(t)=0 when the frequency offset f_(Δ)=0.

In this simplified FOM transmit mixer implementation 1101, circuitcomplexity is reduced by eliminating the Q channel processing circuitsand phase shifters, such as the phase shifters 452, 454 and summingcircuits 456 shown in FIG. 4B. In addition to reducing the requiredhardware, the I-branch only FOM mixer 1101 eliminates the magnitude andphase mismatch problem that can arise from unbalanced I/Q mixing (thatis, misaligned amplitude and phase between I and Q channels). However, adrawback with such single-channel mixer is that the mixer output signalx₁(t) contains an undesirable down-shifted delta component which has tobe properly handled at the receiver.

To illustrate this issue, reference is now made to FIG. 12A whichdepicts a fast-time range FFT spectrum 1200 of a frequency offsetmodulation LFM range division MIMO automotive radar system such as shownin FIG. 3A which uses I-branch only FOM mixers with insufficientfrequency offset in combination with I-channel only analog-to-digitalconverter in the receiver. In this example, the transmitter usesI-channel only FOM mixers connected to all but the first transmitchannel circuit to provide integer multiples of a frequency offsetf_(Δ)=20 Mhz to generate transmit channel offsets at 0 MHz, 20 MHz, 40MHz, 60 MHz, 80 MHz, 100 MHz, 120 MHz, and 140 MHz. At the receiver, theI-channel only ADC having a sampling rate of 320 MHz creates, for eachtransmitter (Txi), a sum component 1201 (shown with the black solid lineand normal font), a delta component 1202 (shown with the black dottedline and bold-faced font), a sum component's complex-conjugate image1203 (shown with the gray solid line and italic font), and a deltacomponent's complex-conjugate image 1204 (shown with the gray dottedline and bold-faced italic font). In an example scenario where a 20 MHzIF spectrum is assumed to be sufficient to cover the maximum range, theI-only FOM mixing results in the delta components whose spectral imagealiases with the sum component, causing significant interference. Inparticular, this is illustrated with the images of the delta component1204 (shown with gray dotted line) for each transmitter (e.g., Tx2)which alias into the designated range spectrum for detecting the sumcomponent 1201 (shown with black solid line) of a different transmitter(e.g., Tx1), thereby causing severe ambiguity in terms of what is beingdetected in the range spectrum segments for each transmit channel. Inaddition, the downshifted delta component 1202 (shown with black dottedline) for each transmitter (e.g., Tx2) aliases into the image of the sumcomponent 1203 (shown with gray solid line) of a different transmitter(e.g., Tx1), thereby impairing correct detection. This happens when thetransmit channel offset is no greater than twice of the instrumentedrange spectrum extent of an individual transmitter's instrumentedrange-spectrum bandwidth.

To provide another illustration of this issue, reference is now made toFIG. 12B which depicts a fast-time range FFT spectrum 1210 of afrequency offset modulation LFM range division MIMO automotive radarsystem such as shown in FIG. 3B which uses I-branch only FOM mixers withinsufficient frequency offset in combination with I-channel onlyanalog-to-digital converter in the receiver. In this example, thetransmitter uses I-channel only FOM mixers connected to each transmitchannel circuit to provide integer multiples of a frequency offsetf_(Δ)=20 Mhz to generate transmit channel offsets at 0 MHz, 20 MHz, 40MHz, 60 MHz, 80 MHz, 100 MHz, 120 MHz, and 140 MHz. At the receiver, theI-channel only ADC having a sampling rate of 320 MHz creates, for eachtransmitter (Txi), a sum component 1211 (shown with the black solid lineand normal font), a delta component 1212 (shown with the dotted blackline and bold-faced font), a sum component's complex-conjugate image1213 (shown with the gray solid line and italic font), and a deltacomponent's complex-conjugate image 1214 (shown with the gray dottedline and bold-faced italic font). As illustrated, the upshifted sumcomponent from the first transmitter Tx1 is cancelled by the downshifteddelta component from the first transmitter Tx1, and the image of the sumcomponent from the first transmitter Tx1 cancels the image of the deltacomponent from the first transmitter Tx1. In addition, the images of thedelta component 1214 (shown with gray dotted line) for one transmitter(e.g., Tx2) alias into the designated range spectrum for detecting thesum component 1211 (shown with black solid line) for a differenttransmitter (e.g., Tx1), thereby causing severe ambiguity in terms ofwhat is being detected in the range spectrum segments for each transmitchannel. Likewise, the downshifted delta component 1212 (shown withblack dotted line) for each transmitter (e.g., Tx2) aliases into theimage of the sum component 1213 (shown with gray solid line) of adifferent transmitter (e.g., Tx1), thereby impairing correct detection.Again, this happens when the transmit channel offset is no greater thantwice of the instrumented range spectrum extent of an individualtransmitter's instrumented range-spectrum bandwidth.

To prevent spectrum aliasing interactions between the images of thedelta component (e.g., 1204) of a first I-channel only mixer and anadjacent channel's sum component (e.g., 1201), the amount of offsetfrequency f_(Δ) should be doubled or increased by at least 100 percent,with a corresponding increase in the ADC sampling rate at the receivermodule 320 in order to maintain number of supported transmitters. As aresult of these adjustments to the amount of frequency offset and ADCsampling rate, the final range spectrum may be derived from the sum anddelta components by means of coherently integrating the range spectrums.By at least doubling the offset frequency f_(Δ), much of the aliasingeffect is eliminated. With the up-shifted sum component and down-shifteddelta component range spectrum both available, they can be combined toachieve better signal-to-noise ratio (SNR).

To illustrate a first example solution, reference is now made to FIG.13A which depicts a fast-time range FFT spectrum 1300 of a frequencyoffset modulation LFM range division MIMO automotive radar system suchas shown in FIG. 3A which uses I-branch only FOM mixer range spectrumwith sufficient frequency offset in combination with I-channel onlyanalog-to-digital converter in the receiver in accordance with selectedembodiments of the present disclosure. In this first example solution,the transmitter uses I-channel only FOM mixers connected to all but thefirst transmit channel circuit to provide integer multiples of afrequency offset f_(Δ)=40 Mhz to generate transmit channel offsets at 0MHz, 40 MHz, 80 MHz, and 120 MHz. At the receiver, the I-channel onlyADC having a sampling rate of 320 MHz creates, for each transmitter(Txi), an upshifted sum component 1301 (shown with the black solid lineand normal font), a downshifted delta component 1302 (shown with thedotted black line and bold-faced font), a sum component'scomplex-conjugate image 1303 (shown with the gray solid line and italicfont), and a delta component's complex-conjugate image 1304 (shown withthe gray dotted line and bold-faced italic font). As illustrated, theupshifted sum component 1301 for a first transmitter (e.g., from thefirst transmitter Tx1) has no interfering aliasing from the image of thedelta component 1304 from an adjacent transmitter (e.g., from the secondtransmitter Tx2). In addition, there is no aliasing interference betweenimages of the delta component 1302 (shown with black dotted line) for afirst transmitter (e.g., a first transmitter Tx1) and the image of thesum component 1303 (shown with gray solid line) for an adjacenttransmitter (e.g., a second transmitter Tx2). As a result, the sum anddelta components of the first transmitter Tx1 are both available and canbe combined in the first IF bandwidth segment (0-20 MHz) to achievebetter SNR performance.

As illustrated with the first example solution shown in FIG. 13A, theADC sampling rate of 320 MHz supports four transmitters, each consuming40 MHz except for the first transmitter Tx1, which consumes only 20 MHz.Thus, the number of supported transmitters is halved in comparison tothe example shown in FIG. 12A, given the same ADC sampling rate (namely320 MHz). However, FIG. 13A shows that there is unused spectrum between140 MHz and 160 MHz and −140 MHz and −160 MHz, indicating that the ADCrate may be reduced by 40 MHz to 280 MHz and still support 4transmitters. Alternatively, the receiver module (e.g., 320) may beconfigured to support additional transmitters by increasing the ADCsampling rate by 80 MHz to fit each additional transmitter (in the givenexample). For example, to fit 8 transmitters, the ADC sampling rateshould be increased to at least 600 MHz. Similarly, to support 16transmitters, the ADC sampling rate must be at least 1240 MHz.

To illustrate a second example solution, reference is now made to FIG.13B which depicts a fast-time range FFT spectrum 1310 of a frequencyoffset modulation LFM range division MIMO automotive radar system suchas shown in FIG. 3A which uses I-branch only FOM mixer range spectrumwith sufficient frequency offset in combination with I-channel onlyanalog-to-digital converter in the receiver in accordance with selectedembodiments of the present disclosure. In this second example, thetransmitter uses I-channel only FOM mixers connected to all transmitchannel circuits to provide integer multiples of a frequency offsetf_(Δ)=40 Mhz to generate transmit channel offsets at 0 MHz, 40 MHz, 80MHz, and 120 MHz. At the receiver, the I-channel only ADC having asampling rate of 320 MHz creates, for each transmitter (Txi), anupshifted sum component 1311 (shown with the black solid line and normalfont), a downshifted delta component 1312 (shown with the dotted blackline and bold-faced font), a sum component's complex-conjugate image1313 (shown with the gray solid line and italic font), and a deltacomponent's complex-conjugate image 1314 (shown with the gray dottedline and bold-faced italic font). As illustrated, if a frequency offsetf_(Δ) is not applied to the first transmitter, the upshifted sumcomponent 1311 for the first transmitter Tx1 is cancelled by the imageof the delta component 1314 for the first transmitter Tx1, and the imageof the delta component 1314 for the first transmitter Tx1 is cancelledby the image of the sum component 1313 for the first transmitter Tx1. Toavoid aliasing and cancellation of the signal from the first transmitterTx1, the first transmitter Tx1 needs to be offset, but only by theregular amount which is at least the instrumented range spectrum extentof a transmitter.

To illustrate an adjustment for avoiding cancellation of the firsttransmitter signal, reference is now made to FIG. 13C which depicts afast-time range FFT spectrum 1320 of a frequency offset modulation LFMrange division MIMO automotive radar system such as shown in FIG. 3Bwhich uses I-branch only FOM mixer range spectrum with a first frequencyoffset for the first transmitter and a second, doubled frequency offsetfor the remaining transmitters in combination with I-channel onlyanalog-to-digital converter in the receiver in accordance with selectedembodiments of the present disclosure. In this example solution, thetransmitter uses I-channel only FOM mixers connected to all transmitchannel circuits to generate transmit channel offsets at 20 MHz, 60 MHz,100 MHz, and 120 MHz. In this example, a first frequency offset (e.g.,f_(Δ)=20 Mhz) is applied to the first transmitter which is at least theinstrumented range spectrum extent of a transmitter, and a secondfrequency offset (e.g., f_(Δ)=40 Mhz) is applied to the remainingtransmitters to avoid signal overlap and cancellation of the firsttransmitter Tx1. At the receiver, the I-channel only ADC having asampling rate of 320 MHz creates, for each transmitter (Txi), anupshifted sum component 1321 (shown with the black solid line and normalfont), a downshifted delta component 1322 (shown with the black red lineand bold-faced font), a sum component's complex-conjugate image 1323(shown with the gray solid line and italic font), and a deltacomponent's complex-conjugate image 1324 (shown with the gray dottedline and bold-faced italic font). As illustrated, the “Tx1” deltacomponent's target echo component 1325 is phase-shifted 180° from the“Tx1” sum component's target echo component 1321. In addition, thecorresponding noise components for the first transmitter (e.g., image ofthe sum component for Tx1 and the image of the delta component for Tx1)are independent of one another. As a result, the delta component of Tx11322 may be phase-shifted by 180° and combined with the sum component ofthe Tx1 1321 to double the target echo component's amplitude, quadrupleits power, thereby doubling the noise component's variance (power) andthe SNR. As will be appreciated, a similar improvement in the SNR gaincan also be achieved by combining the images of the sum and deltacomponents 1323, 1324. However, SNR gains cannot be achieved bycombining a sum or delta component with its image because the noisevalues are not independent.

As illustrated with the example solution shown in FIG. 13C, the ADCsampling rate of 320 MHz supports four transmitters, which is half thenumber of transmitters from the example shown in FIG. 12B given the sameADC sampling rate (namely 320 MHz). However, the receiver module (e.g.,320) may be configured to support additional transmitters by increasingthe ADC sampling rate by 80 MHz to fit each additional transmitter (inthe given example). For example, to fit 8 transmitters, the ADC samplingrate should be increased to at least 600 MHz. Similarly, to support 16transmitters, the ADC sampling rate must be at least 1240 MHz.

To illustrate a first example solution for coherently integrating thesum and delta components, reference is now made to FIG. 14A whichdiagrammatically depicts a coherent integration of the sum and deltacomponents in a fast-time range FFT spectrum 1400 of a frequency offsetmodulation LFM range division MIMO automotive radar system such as shownin FIG. 3A which uses I-branch only FOM mixer range spectrum withsufficient frequency offset in combination with I-channel onlyanalog-to-digital converter in the receiver in accordance with selectedembodiments of the present disclosure. In the depicted example solution,the transmitter uses I-channel only FOM mixers connected to all but thefirst transmit channel circuit to generate transmit channel offsets at 0MHz, 40 MHz, 80 MHz, and 120 MHz. At the receiver, the I-channel onlyADC having a sampling rate of 320 MHz creates, for each transmitter(Txi), an upshifted sum component (shown with the black solid line andnormal font), a downshifted delta component (shown with the dotted blackline and bold-faced font), a sum component's complex-conjugate image(shown with the gray solid line and italic font), and a deltacomponent's complex-conjugate image (shown with the gray dotted line andbold-faced italic font). Since the I-only FOM mixers on transmit moduledivide the transmit power into the sum and delta components, thereceived signal power is also halved if only the sum or delta componentis processed at the receiver (except for the first transmit channel).This is illustrated in FIG. 14A which shows that the transmitter's sumand delta spectral segments are separated by applying the appropriatefrequency offset values to each transmitter channel.

Once the separation of transmitters and the sum and delta spectralsegments are done, a final spectrum can be produced for each transmitterby using a combination of phase shifter and summing circuits 1401-1407to coherently combine the sum and delta spectrums in the digital domain.In this example, it is noted that the first transmitter Tx1 does notundergo the FOM mixing, so it is ahead of the sum component outputs by90 degrees in phase. As a result, a first phase shifter circuit 1401applies a −90° phase shift to the first transmitter's spectrum to alignall outputs (e.g., by multiplying with e^(−jπ/2)) into an output Tx1signal 1410. For the remaining transmitters, it is noted that the sumcomponent for a given transmitter (e.g., Tx2) lags the correspondingdelta component by a phase shift of 180°, so coherent combination isimplemented simply by reversing the sign of the delta component beforesumming the two extracted spectrum segments. For example, a second phaseshifter circuit 1402 applies a 180° phase shift to the secondtransmitter's delta component for combination with the secondtransmitter's sum component at the summing circuit 1403 to generate acoherently combined output Tx2 signal 1411. In addition, a third phaseshifter circuit 1404 applies a 180° phase shift to the thirdtransmitter's delta component for combination with the thirdtransmitter's sum component at the summing circuit 1405 to generate acoherently combined output Tx3 signal 1412. And finally, a fourth phaseshifter circuit 1406 applies a 180° phase shift to the fourthtransmitter's delta component for combination with the fourthtransmitter's sum component at the summing circuit 1407 to generate acoherently combined output Tx4 signal 1413. As a result, the coherentlycombined delta and sum components for the transmitters 1410-1413 resultin a quadrupled target echo signal power, a doubled noise power, and adoubled SNR.

As illustrated with the example solution shown in FIG. 14A, the ADCsampling rate of 320 MHz supports four transmitters. Thus, the number ofsupported transmitters is halved in comparison to the example shown inFIG. 12A, given the same ADC sampling rate (namely 320 MHz). However,FIG. 14A shows that there is unused spectrum between 140 MHz and 160 MHzand between −140 MHz and −160 MHz, indicating that the ADC rate may bereduced by 40 MHz to 280 MHz and still support 4 transmitters.Alternatively, the receiver module may be configured to supportadditional transmitters by increasing the ADC sampling rate to fit eachadditional transmitter.

To illustrate a second example solution for coherently integrating thesum and delta components, reference is now made to FIG. 14B whichdiagrammatically depicts a coherent integration of the sum and deltacomponents in a fast-time range FFT spectrum 1420 of a frequency offsetmodulation LFM range division MIMO automotive radar system such as shownin FIG. 3B which uses I-branch only FOM mixer range spectrum withsufficient frequency offset in combination with I-channel onlyanalog-to-digital converter in the receiver in accordance with selectedembodiments of the present disclosure. In the depicted solution, thetransmitter uses I-channel only FOM mixers connected to all transmitchannel circuits to generate transmit channel offsets at 20 MHz, 60 MHz,100 MHz, and 140 MHz. At the receiver, the I-channel only ADC having asampling rate of 320 MHz creates, for each transmitter (Txi), anupshifted sum component 1421 (shown with the black solid line and normalfont), a downshifted delta component 1422 (shown with the dotted blackline and bold-faced font), a sum component's complex-conjugate image1423 (shown with the gray solid line and italic font), and a deltacomponent's complex-conjugate image 1424 (shown with the gray dottedline and bold-faced italic font). Since the I-only FOM mixers ontransmit module divide the transmit power into the sum and deltacomponents, the received signal power is also halved if only the sum ordelta component is processed at the receiver. This is illustrated inFIG. 14B which shows that the transmitter's the sum and delta spectralsegments are separated by applying the appropriate frequency offsetvalues to each transmitter channel.

Once the separation of transmitters and the sum and delta spectralsegments are done, a final spectrum can be produced for each transmitterby using a combination of phase shifter and summing circuits 1425-1432to coherently combine the sum and delta spectrums in the digital domain.In this example, each of the transmitters generates a sum component thatlags the corresponding delta component by a phase shift of 180°, socoherent combination is implemented simply by reversing the sign of thedelta component before summing the two extracted spectrum segments. Forexample, a first phase shifter circuit 1425 applies a 180° phase shiftto the first transmitter's delta component for combination with thefirst transmitter's sum component at the summing circuit 1426 togenerate a coherently combined output Tx2 signal 1432. In addition, asecond phase shifter circuit 1427 applies a 180° phase shift to thesecond transmitter's delta component for combination with the secondtransmitter's sum component at the summing circuit 1428 to generate acoherently combined output Tx2 signal 1433. In addition, a third phaseshifter circuit 1429 applies a 180° phase shift to the thirdtransmitter's delta component for combination with the thirdtransmitter's sum component at the summing circuit 1430 to generate acoherently combined output Tx3 signal 1434. And finally, a fourth phaseshifter circuit 1431 applies a 180° phase shift to the fourthtransmitter's delta component for combination with the fourthtransmitter's sum component at the summing circuit 1432 to generate acoherently combined output Tx4 signal 1435. As a result, the coherentlycombined delta and sum components for the transmitters 1432-1435 resultin a quadrupled target echo signal power, a doubled noise power, and adoubled SNR.

As illustrated with the example solution shown in FIG. 14B, the ADCsampling rate of 320 MHz supports four transmitters. Thus, the number ofsupported transmitters is halved in comparison to the example shown inFIG. 12B, given the same ADC sampling rate (namely 320 MHz). However,the receiver module may be configured to support additional transmittersby increasing the ADC sampling rate to fit each additional transmitter.

To illustrate a third example solution for coherently integrating thesum and delta image components, reference is now made to FIG. 15 whichdiagrammatically depicts a coherent integration of the sum and deltaimage components in a fast-time range FFT spectrum 1500 of a frequencyoffset modulation LFM range division MIMO automotive radar system suchas shown in FIG. 3B which uses I-branch only FOM mixer range spectrumwith sufficient frequency offset in combination with I-channel onlyanalog-to-digital converter in the receiver in accordance with selectedembodiments of the present disclosure. In the depicted solution, thetransmitter uses I-channel only FOM mixers connected to all transmitchannel circuits to generate transmit channel offsets at 20 MHz, 60 MHz,100 MHz, and 140 MHz. At the receiver, the I-channel only ADC having asampling rate of 320 MHz creates, for each transmitter (Txi), anupshifted sum component 1501 (shown with the black solid line and normalfont), a downshifted delta component 1502 (shown with the dotted blackline and bold-faced font), a sum component's complex-conjugate image1503 (shown with the gray solid line and italic font), and a deltacomponent's complex-conjugate image 1504 (shown with the gray dottedline and bold-faced italic font). Since the I-only FOM mixers ontransmit module divide the transmit power into the sum and deltacomponents, the received signal power is also halved if only the sum ordelta component is processed at the receiver. This is illustrated inFIG. 15 which shows that the transmitter's the sum and delta spectralsegments are separated by applying the appropriate frequency offsetvalues to each transmitter channel.

Once the separation of transmitters and the sum and delta image spectralsegments are done, a final spectrum can be produced for each transmitterby using a combination of phase shifter, summing, and index reversalcircuits 1505-1516 to coherently combine the sum and delta imagespectrums in the digital domain. In this example, each of thetransmitters generates a sum image component that lags the correspondingdelta image component by a phase shift of 180°, so coherent combinationis implemented simply by reversing the sign of the sum image componentsegment before summing with the delta image component segment. Forexample, a first phase shifter circuit 1505 applies a 180° phase shiftto the first transmitter's sum image component for combination with thefirst transmitter's delta image component at the summing circuit 1506 togenerate a coherently combined signal which has sample indices reversedat the index reversal circuit 1513 to generate the output Tx2 signal1520. In addition, a second phase shifter circuit 1507 applies a 180°phase shift to the second transmitter's sum image component forcombination with the second transmitter's delta image component at thesumming circuit 1508 to generate a coherently combined signal which hassample indices reversed at the index reversal circuit 1514 to generatethe output Tx2 signal 1521. In addition, a third phase shifter circuit1509 applies a 180° phase shift to the third transmitter's sum imagecomponent for combination with the third transmitter's delta imagecomponent at the summing circuit 1510 to generate a coherently combinedsignal which has sample indices reversed at the index reversal circuit1515 to generate the output Tx3 signal 1522. And finally, a fourth phaseshifter circuit 1511 applies a 180° phase shift to the fourthtransmitter's sum image component for combination with the fourthtransmitter's delta image component at the summing circuit 1512 togenerate a coherently combined signal which has sample indices reversedat the index reversal circuit 1516 to generate the output Tx4 signal1523. As a result, the coherently combined delta and sum imagecomponents for the transmitters 1520-1523 result in a quadrupled targetecho signal power, a doubled noise power, and a doubled SNR.

As illustrated with the example solution shown in FIG. 15 , the ADCsampling rate of 320 MHz supports four transmitters. Thus, the number ofsupported transmitters is halved in comparison to the example shown inFIG. 12B, given the same ADC sampling rate (namely 320 MHz). However,the receiver module may be configured to support additional transmittersby increasing the ADC sampling rate to fit each additional transmitter.

To provide additional details for an improved understanding of selectedembodiments of the present disclosure, reference is now made to FIG. 16Awhich depicts a fast-time range FFT spectrum 1600 of a frequency offsetmodulation LFM range division MIMO automotive radar system such as shownin FIG. 3A which uses I-branch only FOM mixer range spectrum withsufficient frequency offset in combination with I/Q-channelanalog-to-digital converter in the receiver. In the depicted embodiment,the transmitter uses I-channel only FOM mixers connected to all but thefirst transmit channel circuit to generate transmit channel offsets at 0MHz, 20 MHz, 40 MHz, 60 MHz, 80 MHz, 100 MHz, 120 MHz, and 140 MHz. Atthe receiver, an I/Q channel ADC is provided with a sampling rate of 320MHz to create, for each transmitter (Txi), an upshifted sum component1601 (shown with the black solid line and normal font) and a downshifteddelta component 1602 (shown with the dotted black line and bold-facedfont). Since the I-only FOM mixers on transmit module divide thetransmit power into the sum and delta components, the received signalpower is also halved if only the sum or delta component is processed atthe receiver (except for the first transmit channel). When I-channelonly FOM mixers are used at the transmitter and the I/Q ADC is used atthe receiver, the delta components may alias into sum components ifsampling frequency is not sufficiently high, causing cancelation orinterference. FIG. 16A shows the case of the minimally required samplingfrequency for the I/Q ADC.

Once the separation of transmitters and the sum and delta spectralsegments are done, a final spectrum can be produced for each transmitterby using a combination of phase shifter and summing circuits tocoherently combine the sum and delta spectrums in the digital domain. Inthis example, it is noted that the first transmitter Tx1 does notundergo the FOM mixing, so it is ahead of the sum component outputs by90 degrees in phase. As a result, the sum and delta components of thetransmitters Tx2-Tx8 can be coherently combined after applying the 180°phase shift to the delta component, while the sum component of the firsttransmitter Tx1 is phase shifted by −90° to be in phase with otherchannels. As illustrated with the example solution shown in FIG. 16A,the ADC sampling rate of 320 MHz supports eight transmitters.

To provide additional details for an improved understanding of selectedembodiments of the present disclosure, reference is now made to FIG. 16Bwhich depicts a fast-time range FFT spectrum 1610 of a frequency offsetmodulation LFM range division MIMO automotive radar system such as shownin FIG. 3B which uses I-branch only FOM mixer range spectrum withinsufficient frequency offset in combination with I/Q-channelanalog-to-digital converter in the receiver. In the depicted embodiment,the transmitter uses I-channel only FOM mixers connected to all but thefirst transmit channel circuit to generate transmit channel offsets at 0MHz, 20 MHz, 40 MHz, 60 MHz, 80 MHz, 100 MHz, 120 MHz, and 140 MHz. Atthe receiver, an I/Q channel ADC is provided with a sampling rate of 320MHz to create, for each transmitter (Txi), an upshifted sum component1611 (shown with the black solid line and normal font) and a downshifteddelta component 1612 (shown with the dotted black line and bold-facedfont). Since the I-only FOM mixers on transmit module divide thetransmit power into the sum and delta components, the received signalpower is also halved if only the sum or delta component is processed atthe receiver (except for the first transmit channel). When I-channelonly FOM mixers are used at the transmitter and the I/Q ADC is used atthe receiver, the delta components may alias into sum components ifsampling frequency is not sufficiently high, causing cancelation orinterference. FIG. 16B shows the case where the upshifted sum componentfrom the first transmitter Tx1 is cancelled by the downshifted deltacomponent from the first transmitter Tx1.

To avoid interference or cancellation of the first transmitter Tx1, acarrier frequency offset should also be applied to the first transmitterTx1 and the sampling frequency should be increased to fit alltransmitters unambiguously. An example embodiment of such a solution isillustrated in FIG. 16C which depicts a fast-time range FFT spectrum1620 of a frequency offset modulation LFM range division MIMO automotiveradar system such as shown in FIG. 3B which uses I-branch only FOM mixerrange spectrum with sufficient frequency offset in combination withI/Q-channel analog-to-digital converter in the receiver in accordancewith selected embodiments of the present disclosure. In the depictedembodiment, the transmitter uses I-channel only FOM mixers connected toall transmit channel circuits to generate transmit channel offsets at 20MHz, 40 MHz, 60 MHz, 80 MHz, 100 MHz, 120 MHz, 140 MHz, and 160 MHz. Atthe receiver, an I/Q channel ADC is provided with a sampling rate of 320MHz to create, for each transmitter (Txi), an upshifted sum component1611 (shown with the black solid line and normal font) and a downshifteddelta component 1612 (shown with the dotted black line and bold-facedfont). Since the I-only FOM mixers on transmit module divide thetransmit power into the sum and delta components, the received signalpower is also halved if only the sum or delta component is processed atthe receiver (except for the first transmit channel). When I-channelonly FOM mixers at the transmitter (including the first transmitter Tx1)are each offset by an integer multiple of the frequency offset f_(Δ) andthe I/Q ADC at the receiver uses a sufficiently high sampling frequency,the delta components do not alias into the sum components, therebyavoiding cancelation or interference.

Once the separation of transmitters and the sum and delta spectralsegments are done, a final spectrum can be produced for each transmitterby using a combination of phase shifter and summing circuits tocoherently combine the sum and delta spectrums in the digital domain,provided that the ADC sampling frequency is increased to fit alltransmitters unambiguously. In this example, it is noted that the firsttransmitter Tx1 undergoes FOM mixing along with the rest of thetransmitters. As a result, the sum and delta components of thetransmitters Tx1-Tx8 can be coherently combined after applying the 180°phase shift to the respective delta components. As illustrated with theexample solution shown in FIG. 16C, the ADC sampling rate of 360 MHzsupports eight transmitters.

While selected embodiments of the FOM MIMO scheme are described withreference to using an offset frequency generator to generate uniquefrequency offset tones for each transmit channel, it will be appreciatedthat other frequency offset modulation schemes may also be used. Forexample, selected FOM implementations may use a fast-time phase shifterat each transmit channel circuit of the transmit module in combinationwith a high speed ADC at the receiver module, thereby eliminating theneed for I/Q modulation mixers at the transmit module.

To provide additional details for an improved understanding of selectedphase shifter embodiments of the present disclosure, reference is nowmade to FIG. 17 which depicts a simplified schematic block diagram of afrequency offset modulation LFM range division MIMO automotive radarsystem 1700 which includes an LFM RD-MIMO radar device 1730 havingfast-time phase shifters in the transmit module 1710 which are connectedand configured to transmit and receive LFM waveforms 1702, 1703 forreflection by a target 1701 to the receive module 1720 under control ofa radar controller processor (not shown). In selected embodiments, theLFM RD-MIMO radar device 1730 and/or radar controller processor may beembodied as a line-replaceable unit (LRU) or modular component that isdesigned to be replaced quickly at an operating location. In additionand as described hereinbelow, the LFM RD-MIMO radar device 1730 may alsobe configured to perform time-division multiplexing of the transmittedLFM waveforms 1702, 1703 to implement a combined time-division andrange-division MIMO scheme to separate the transmitters not only in therange domain, but also in the time domain.

As depicted, each radar device 1730 includes one or more transmittingantenna elements TXi and at least a first receiving antenna element RXconnected, respectively, to one or more radio frequency (RF) transmitmodules 1710 and receive module 1720. At each transmit module 1710, atransmit channel circuit is provided for each transmit antenna. Forexample, a first transmit channel circuit includes a first RFconditioning module 1711 and power amplifier 1712 connected to a firsttransmit antenna TX₁, a second transmit channel circuit includes asecond RF conditioning module 1714 and power amplifier 1715 connected toa second transmit antenna TX₂, and so on with the Nth transmit channelcircuit including an Nth RF conditioning module 1717 and power amplifier1718 connected to the Nth transmit antenna TX_(N).

In addition, each radar device 1730 includes a chirp generator 1704which is configured and connected to supply a chirp input signal 1705 tothe different transmit channel circuits 1711/1712, 1714/1715, 1717/1718in the transmitter module(s) 1710. However, instead of providing thechirp input signal 1705 directly to all of the transmit channelcircuits, the radar device 1730 also includes a programmable fast-timephase shifter circuits which are connected to phase shift the chirpinput signal 1705, thereby creating progressive phase shifts which mimicthe effects of a frequency offset for the transmit antennas(TX₁-TX_(N)). To this end, the first transmit channel circuit 1711/1712may be connected to directly receive the chirp input signal 1705.However, the second transmit channel circuit 1714/1715 may include afirst fast-time phase shifter 1713 which is connected to apply a firstphase shift to the chirp input signal 1705 before being filtered andamplified by the second transmit channel circuit 1714/1715 fortransmission over the antenna TX₂. In similar fashion, the remainingtransmit channel circuits (e.g., 1717/1718) may include a fast-timephase shifter (e.g., 1716) which is connected to apply a unique phaseshift to the chirp input signal 1705 before being filtered and amplifiedby the antenna (e.g., TX_(N)).

As disclosed herein, each of the phase shifters 1713, 1717 may beimplemented with a programable phase shifter having a fast switchingtime or response time. For example, phase shift control signals 1706 canbe applied to programmable K-bit phase shifters 1713, 1716 to createprogressive phase shifts at the transmitter module 1710, therebymimicking the effects of frequency offset for 2^((K-1)) transmitters. Inselected embodiments, the phase shifters 1713, 1716 are used tointroduce regular progressive phase shift at fast-time samplingintervals. For example, a 1-bit phase shifter with switching positions{0°, 180° } can cause frequency offsets of {0,1/(2T_(s))} [Hz], whereT_(s) is fast-time sampling interval as well as the switching intervalof the phase shifter in seconds. For another example, a 4-bit phaseshifter with switching positions {0°, 22.5°, 45°, 67.5°, 90°, 112.5°,135°, 157.5°, 180°, 202.5°, 225°, 247.5°, 270°, 292.5°, 315°, 337.5° }can cause frequency offsets of {0, 1/(32T_(s)), 2/(32T_(s)),3/(32T_(s)), 4/(32T_(s)), 5/(32T_(s)), 6/(32T_(s)), 7/(32T_(s))} [Hz].In this case, a T_(s) corresponding to a switching frequency of 320 MHzallows for 8 transmitters to share a total of 320 MHz of fast-timespectrum, each occupying a segment of 40 MHz, whose integer {0, 1, 2 . .. 7} multiples are the amount of frequency offsets applied to thetransmitters.

In principle, a K-bit phase shifter can support up to 2^((K-1))transmitters. In addition, a frequency offset of f_(Δ) Hz is equivalentto imposing a progressive phase shift in time by a rate of 2πf_(Δ)radians per second. At a sampling rate of f_(Δ) Hz (or sampling intervalof T_(s) which equals to 1/f_(s)), the amount of progressive (oradditive) phase shift per T_(s) interval is then 2πf_(Δ)/f_(s) radians.To comply with the Nyquist sampling theorem, the phase shifter switchedat fast-time sampling rate can support a maximum frequency offset of1/(2T_(s)) Hz.

In one example, the transmitter module 1710 may be programmed with phaseshift control signals 1716 so that the phase shifters 1713, 1716 providea progressive phase shift of 45 degrees (e.g., {0°, 45°, 90°, 135°,180°, 225°, 270°, 315°, 0°, 45°, . . . }) at intervals of T_(sw)seconds. With this arrangement, a complete a phase shift is imposed onthe reference chirp every 8 intervals such that an effective frequencyoffset of ⅛T_(s) Hz is imposed on the chirp signal. For another example,the transmitter module 1710 may be programmed with a 337.5° progressphase shift per switching interval, resulting in the application of thefollowing progressive phase shift: {0°, 337.5°, 315°, 292.5°,270°,247.5°,225°, 202.5°, 180°, 157.5°, 135°, 112.5°, 90°, 67.5°, 45°,22.5°,0°, . . . }. The resulting progressive phase shift has a step sizeof −22.5° to effectively apply a negative frequency offset. Based onNyquist criteria, a progressive phase shift with a step size no morethan 180° may be applied without incurring ambiguity. As a result, 4-bitphase shifters can be used to support up to 8 frequency offsets(including the zero offset).

To ensure sufficient range space on the transmit spectrum is availablefor division amongst all transmitters, a faster analog-to-digitalconverter (ADC) 1726 may be employed at each receive channel 1720. Forexample, an LFM RD-MIMO radar device 1730 using fast-time phase shifters1713, 1716 to implement frequency offset modulation should use an ADCsampling rate that is increased to N×40 Msps. As a result, the fast-timeFFT processing can divide the spectrum into N consecutive segments, witheach being associated with a corresponding transmitter.

With the fast-time phase shifters 1713, 1716 phase-shifting the chirpinput signal 1705 under control of the phase shift control signals 1706before transmission on the transmit antennas TX₁-TX_(N), the combinedreflected LFM waveforms 1703 are received and processed by the receivermodule 1720. In particular, the receive antenna RX receives the combinedreflected LFM waveforms 1703 which are then amplified by the low noiseamplifier (LNA) 1721. At the I/Q mixer 1722, the amplified receivesignal is mixed with the reference chirp signal 1705 before beingconditioned for digital conversion by the high pass filter 1723,variable gain amplifier 1724, low pass filter 1725, andanalog-to-digital converter 1726.

Generally speaking, the implementation of frequency offset modulationusing fast-time phase shifters differs from a phase-coded chirp systemwhere the phase shift switching interval may not coincide with thefast-time sampling rate and where the phase shift usually follows anorthogonal code pattern which is not a progressive phase shift. Incode-division MIMO systems, the transmitters are separated bytransmitting a unique phase-coded waveform that is orthogonal to thoseof other transmitters, and the receivers require a receiver bank ofcorrelators to decode the signals from individual transmitters. Thisdiffers from the range-division principle of the present invention.

As disclosed herein, the use of frequency offset modulation in an LFMautomotive radar systems enables very large MIMO arrays to be formed byseparating transmitter signals in the fast-time Fourier or the rangedomain. The resulting virtual array is much larger than the conventionalTD-MIMO approach, thereby achieving high angular resolution performance.However, by combining the FOM RD MIMO approach with TD MIMO approach,even larger MIMO virtual arrays can be formed with the additionalbenefit of mitigating or reducing the problem of strongbeyond-maximum-range target interference than an arise with FOM RD MIMOonly systems. In particular, this combined approach allows theseparation of the transmitters not only in the range domain, but also inthe time domain to allow the detection of extremely large RCS targetsthat are beyond the maximum unambiguous range measurable for eachtransmitter. To implement the time-domain separation, the LFM RD-MIMOradar device 330A may be implemented as a combination LFMRD-MIMO/TD-MIMO radar device 330A which is configured to activateadjacent transmitters in an alternating fashion over time. With thetime-domain modulation implemented at the transmit module 310A, thereceived transmit spectrum includes, for each transmitter range spectrumsegment, an adjacent range spectrum segment that is vacant, therebyallowing strong beyond-the-range targets to be freely present andcorrectly detected.

One of the benefits from combining the FOM and TD MIMO approaches arisesfrom the detection distance limitations of the FOM MIMO approach alone.In particular, the FOM MIMO approach effectively divides, at eachreceive channel, the entire range spectrum into N segments, eachcontaining the range spectrum of a corresponding individual transmitter.In some cases, the maximum range extent of an individual range spectrumis not large enough to contain all detectable targets (especially fortargets with extremely large radar cross-sections) at a distance that islonger than the maximum range of the individual transmitter. In suchcases, these strong beyond-the-range targets show up at the rangespectrum of the next transmitter, causing false detections or at leastambiguous detections. To avoid this, a longer-range extent may beallocated to each transmitter by allocating more IF frequency spectrumfor each transmitter. However, this results in higher ADC sampling raterequirements or fewer transmitters fitting into the total range spectrumfor the same ADC sampling rate. This trade-off can be avoided bycombining the time division MIMO principle with the FOM Range DivisionMIMO principle to allow larger range extent while not reducing themaximum number of transmitters that can be supported at the expense ofincreased frame duration.

As disclosed herein, the TD and RD MIMO approaches can be combined witha variety of different schemes. For example, a combined TD and RD MIMOscheme may separately transmit the odd-number transmitters and theeven-number transmitters in two groups so that the transmission for eachchirp is divided with the first group of odd-numbered transmitters beingtransmitted first, and with the second group of even-numberedtransmitters being transmitted second. To illustrate an examplegrouping, reference is now made to FIG. 18 which depicts first andsecond fast-time range FFT spectrums 1801, 1802 for two time slots of areceiver channel of an I/Q sample for an 8-transmitter MIMO automotiveradar which employs both time-division and frequency offset modulatedLFM range-division radar techniques. In this example, it is assumed thatthe range spectrum is divided by transmitters Tx1, Tx2, Tx3, TxN in asequential fashion and adjacent-numbered transmitters' range spectrumare also adjacent to each other. In the depicted arrangement, the firstgroup is first transmitted from the odd-numbered transmitters (e.g.,Tx1, Tx3, Tx5, Tx7), followed by transmission from the even-numberedtransmitters (e.g., Tx2, Tx4, Tx6 Tx8). In the range spectrum of timeslot #1 1801, the even-numbered transmitters do not transmit so thecorresponding range spectrum segments Tx2, Tx4, Tx6, Tx8 are vacant,allowing strong beyond-the-range targets to be present and detected inthese vacant segments in an unambiguous fashion. Likewise, in the rangespectrum of time slot #2 1802, the odd-numbered transmitters Tx1, Tx3,Tx5, Tx7 do not transmit which leaves the corresponding range spectrumsegments vacant so that strong beyond-the-range targets can be freelypresent and detected. This approach effectively doubles the maximumrange of the radar system. As seen from this example, the temporalseparation of transmissions from the first group from the second groupin the time domain, in combination with separation in the frequency orrange domain, greatly reduces the chance of a strong beyond-the-rangetarget interference between adjacent transmitters. While the combined TDand RD approach greatly reduces the risk of interference due to strongbeyond-the-range targets, if such interference is not a practicalconcern, the combined TD and RD approach also increases the number oftransmitters supported.

As will be noted, the last transmitter (e.g., Tx8) does not haveadditional free room for ambiguous detection of the strongbeyond-the-range targets due to the conjugate symmetric nature of theI-sample only spectrum. Such ambiguity may be tolerated because it doesnot occur in the rest of transmitters' range spectrums. Because of suchinconsistency in the range spectrum, the ambiguous target will not becoherently integrated in the subsequent Doppler and angle processingsuch that its impact is minimized. If such tolerance is not acceptable,the limitation may be addressed by increasing the ADC sampling rate byone transmitter's IF frequency extent. Alternatively, the maximum numberof transmitters should be reduced by one without increased ADC samplingrate. In yet another alternative, the number of time slots can beincreased (e.g., more than 2) to further reduce the chance of ambiguousdetections. In such embodiments, the activated transmit antennas in anyparticular transmission time slot are separated from one another by twoor more deactivated transmit antennas, depending on the number of timeslots.

As disclosed herein, the FOM MIMO approach can be combined with otherMIMO approaches besides TD MIMO schemes. For example, a combined DopplerDivision and Range Division MIMO can be implemented to separate thetransmitters in both the range domain (using FOM MIMO) and in theDoppler domain. Likewise, a combined Time Division, Doppler Division,and Range Division MIMO can be implemented to separate the transmittersnot only in the range domain, but also in the Time and Doppler domains.

To provide additional details for an improved understanding of selectedembodiments of the present disclosure, reference is now made to FIG. 19which depicts a simplified flow chart 1900 showing the logic for usingfrequency offset modulation techniques to form virtually large MIMOradar arrays. In an example embodiment, the control logic andmethodology shown in FIG. 19 may be implemented as hardware and/orsoftware on a host computing system, processor, or microcontroller unitthat includes processor and memory for storing programming control codefor constructing and operating a large virtual MIMO radar arrays byintroducing frequency offset modulations signals to reference chirpsignals to enable separation of the transmitter signals in the fast-timeFourier or the range domain.

The process starts (step 1901), such as when the radar system begins theprocess of sensing the location and movement of one or more targetobjects using one or more transmit radar signals that are sent over aplurality of transmit antennas. To generate the transmit radar signals,the radar system first generates a reference chirp signal (step 1902),such as by periodically modulating a transmit radar signal with afrequency and/or phase shift. For example, with automotive FrequencyModulation Continuous Wave (FMCW) radars, the reference chirp signal maybe generated as a Linear Frequency Modulation (LFM) waveform that isdistributed to a plurality of transmit channel circuits which arerespectively associated with a plurality of transmit antennas.

At step 1904, the reference chirp signal is applied to a plurality offrequency offset modulation (FOM) mixers to generate frequency offsetchirp signals for a plurality of transmit channels. In selectedembodiments, the FOM mixing step may be implemented by applying thereference chirp signal to a plurality of FOM mixers which are eachrespectively connected to receive a plurality of defined frequencyoffset tones for mixing with the reference chirp signal, therebygenerating a plurality of different frequency offset reference chirpsignals for use with the plurality of transmit channel circuits. Inaddition, one of the transmit channel circuits may be connected todirectly receive the reference chirp signal without any frequency offsetmodulation. In selected embodiments, the frequency offset mixer may beimplemented with an I/Q channel modulation mixer, an I-channel onlymodulation mixer, or a Q-channel only modulation mixer to implement aspectrum-coherent integration approach. By using frequency offsetmodulation mixers to mix the reference chirp signal as an LFM waveformat each transmit channel with different frequency offset signals (e.g.,Δf, 2Δf, etc.), the receiver may employ a high sampling rate ADC toallow separation of different transmitters' transmit signals in thereceived range spectrum.

As an alternative step 1905, the reference chirp signal is applied to aplurality of fast-time phase shifters to generate phase-shifted chirpsignals for a plurality of transmit channels. In selected embodiments,the phase shifting step may be implemented by applying the referencechirp signal to a plurality of phase shifters which are respectivelycontrolled by a phase shift control signal, thereby generating aplurality of different frequency offset reference chirp signals for usewith the plurality of transmit channel circuits. In addition, one of thetransmit channel circuits may be connected to directly receive thereference chirp signal without any phase shift modulation. By usingphase shifters to introduce regular progressive phase shifts to thereference chirp signal at each transmit channel example, the phaseshifters effectively mimic the effects of frequency offset modulation,thereby enabling the receiver to employ a high sampling rate ADC toallow separation of different transmitters' transmit signals in thereceived range spectrum.

As an optional step 1906, time division modulation may be applied to theplurality of different frequency offset reference chirp signals for usewith the plurality of transmit channel circuits. As indicated with thedashed lines, the time division modulation step may be omitted orskipped in the disclosed sequence 1900. However, selected embodiments ofthe time division modulation step may employ an alternating transmissionscheme whereby multiple transmit time slots are defined such that afirst set of alternating transmit channels (e.g., even-numberedtransmitters) are active in a first time slot and are suppressed in asecond time slot, while a second set of alternating transmit channels(e.g., odd-numbered transmitters) are active in a second time slot andare suppressed in a first time slot. In this configuration, for eachtransmitter, its adjacent range spectrum segment is vacant, therebyenabling strong beyond-the-range targets to be corrected detectedwithout imposing target interference.

At step 1908, the frequency offset or phase-shifted reference chirpsignals are conditioned and amplified for transmission over thecorresponding transmit channel circuits. In selected embodiments, thisprocessing is performed by the transmit channel circuits which eachinclude an RF conditioning module (which filters the output of thecorresponding FOM mixer or phase shifter) and power amplifier (whichamplifies the RF conditioning module output for transmission over acorresponding transmit antenna). In embodiments where time-domainmodulation is used in combination with the frequency/phase offsetmodulation, the non-adjacent transmit channel circuits may be controlledto sequentially condition and amplify transmit radar waveforms fromnon-adjacent transmit antennas.

At step 1910, the reflected frequency/phase offset reference chirpsignals from the different transmit channels are received and amplifiedat the receiver. In selected embodiments, one or more receive antennasat the receiver module receive target returns from the transmittedfrequency/phase offset reference chirp signal waveforms as (radiofrequency) antenna signals for subsequent amplification, such as byusing a low noise amplifier to generate an amplified RF signal from thetarget returns.

At step 1912, the amplified transmit channel signals are mixed with thereference chirp signal at the receiver to generate an intermediatefrequency (IF) signal. In selected embodiments, the mixing step may beimplemented by applying the reference chirp signal to a receiver modulemixer which is also connected to receive the amplified transmit channelsignals for mixing with the reference chirp signal, thereby generatingan intermediate frequency signal.

At step 1914, the intermediate frequency signal is conditioned fordigital conversion. In selected embodiments, the conditioning processincludes feeding the intermediate frequency signal to a high-passfilter, amplifying the filtered signal with a variable gain amplifierbefore being fed to a low-pass filter, thereby generating a re-filteredsignal.

At step 1916, the re-filtered conditioned IF signal is fed to ahigh-speed analog/digital converter (ADC) which has a digital signaloutput that is suitable for digital processing. Because the maximumunambiguous range extent for each frequency offset reference chirpsignal is inversely related to the fast-time sampling interval, the ADChas a high sampling rate. For example, if a conventional TD-MIMO FCMradar uses a 40 mega-samples-per-second (Msps) ADC in the receivermodule, the ADC sampling rate is increased to N×40 Msps to enable theN-transmitters MIMO operation using the disclosed FOM approach. Notealso that depending on the radar system requirements and also upon theactual performance of the low-pass filter that directly precedes theADC, the sample rate of the ADC may need to be increased beyond N×40Msps.

At step 1918, the digital processing is applied to separate thereflected transmit channel signals in the fast-time FFT or range domain,along with other radar signal processing steps. While any suitable radarsignal processing steps may be used, each radar may be configured toperform fast-time FFT and slow-time FFT processing on the received radarsignal to derive range and Doppler information. In the fast-time FFTprocessing, the frequency offset modulation of the reference chirpsignals sent over the N transmission channels enables the spectrum to bedivided into N consecutive segments with each being associated with acorresponding transmitter. Because the transmitters are separated ordivided in the range domain and the waveform is based on LFM, theapproach can also be referred to as the LFM range-division (RD) MIMOapproach. Based on the range-spectrum division arrangement, thefast-time samples associated with distinct transmitters are thenrecovered (and whose sum and delta components are coherently summed forthe case of I-channel only FOM,) and the subsequent MIMO virtual arrayprocessing can be carried out.

At step 1920, the virtual MIMO array is constructed from the reflectedtransmit channel signals which originated from distinct transmitchannels. In selected embodiments, the frequency/phase offset referencechirp signal target return data samples received from the distincttransmit channels are processed using mono-static and/or bi-static radarprinciples to construct and accumulate MIMO virtual array outputs.

At step 1922, the MIMO virtual array outputs are processed by range,Doppler, and angle estimation processes and the target map is generatedto identify the range, Doppler, and angle values for each detectedtarget. The range, Doppler, and angle estimators are typically based onFast Fourier Transform (FFT) and Discrete Fourier Transform (DFT)processors. More advanced spectral estimators including but not limitedto Multiple-signal Classifier (MUSIC) and Estimator of Signal Parametersvia Rotational Invariance Technique (ESPRIT) processors, may also beused for angle processing. In selected embodiments, the radar controllerprocessor may be configured to produce map data identifying paired range(r), Doppler ({dot over (r)}) and angle (θ) values for eachdetected/target object.

As disclosed herein, selected embodiments of the disclosed frequencyoffset modulation range division MIMO radar system may provide severalenhancements when compared with conventional radar systems. In additionto enabling the construction of very large MIMO arrays for automotiveFrequency Modulation Continuous Wave (FMCW) radars that transmit LinearFrequency Modulation (LFM) waveforms, the disclosed radar system can useRF front-end and signal processing blocks of existing radar designswithout significant modifications, thereby minimizing the cost ofdeveloping the new solution. In addition, the combination of FOM RD andTD MIMO approaches enables strong beyond-the-range targets to becorrected detected without imposing target interference. In addition,the present disclosure enables the number of virtual antenna elements tobe constructed via a MIMO approach to equal the product of the number ofphysical transmit and receiver antenna elements, thereby forming alarger aperture than can be formed from the total number of physicalelements and improving the angular resolution.

By now it should be appreciated that there has been provided a radararchitecture, circuit, method, and system in which a reference signalgenerator is configured to produce a transmit reference signal asequence of waveforms (e.g., a chirp signal). In addition, a waveformgenerator is configured to produce a plurality of transmit signals, eachhaving a different frequency offset from the transmit reference signal.In selected embodiments, the waveform generator includes a frequencyoffset signal generator for generating a plurality of differentfrequency offset tones, and a plurality of frequency offset modulationmixers, each connected to mix the transmit reference signal with one ofthe plurality of different frequency offset tones, thereby generating afrequency offset transmit reference signal as one of the N radiofrequency encoded transmit signals. In other embodiments, the waveformgenerator includes a plurality of fast-time phase shifters, eachconnected to receive the transmit reference signal, thereby generating afrequency offset transmit reference signal as one of the N radiofrequency encoded transmit signals. In such embodiments, the pluralityof fast-time phase shifters may include a bank of K-bit phase shiftersfor introducing a regular progressive phase shift to the transmitreference signal at fast-time sampling intervals. The radar system alsoincludes a signal encoder to encode the plurality of transmit signalsusing a signal conditioning and power amplification to produce andtransmit N radio frequency encoded transmit signals over N transmitantennas. In selected embodiments, the signal encoder is configured toencode the plurality of transmit signals using time divisionmultiplexing to produce a first set of N radio frequency encodedtransmit signals for transmission over a first set of N transmitantennas during a first time slot, and to produce a second set of Nradio frequency encoded transmit signals for transmission over a secondset of N transmit antennas during a second time slot. In suchembodiments, the signal encoder may be configured activate even-numberedtransmit antennas for transmitting the first set of N radio frequencyencoded transmit signals during the first time slot, and to activateodd-numbered transmit antennas for transmitting the second set of Nradio frequency encoded transmit signals during the second time slot. Ata receiver module, at least a first receive antenna is provided toreceive a target return signal reflected from the N radio frequencyencoded transmit signals by a target. The receiver module also includesa downconverter that is configured to mix the target return signal withthe transmit reference signal, thereby producing an intermediatefrequency signal. In addition, a high-speed analog-to-digital converteris connected to convert the intermediate frequency signal to a digitalsignal. In selected embodiments, a notch filter bank is connectedbetween the downconverter and the high-speed analog-to-digital converterfor filtering the intermediate frequency signal to remove zero-rangeinterference in a fast-time spectrum of each transmitter. In otherembodiments, the downconverter is an I/Q mixer which is connected tofirst and second high-speed analog-to-digital converters whichrespectively convert the intermediate frequency signal to an I-channeldigital signal and Q-channel digital signal. In selected FOMembodiments, the high-speed analog-to-digital converter has a samplingrate of at least N times of the minimally required sampling rate of asingle-transmitter system that detects targets unambiguously in range.In selected TD/FOM embodiments, the high-speed analog-to-digitalconverter has a sampling rate of at least 2*N times of the minimallyrequired sampling rate of a single-transmitter system that detectstargets unambiguously in range. The radar system also includes a radarcontrol processing unit that is configured to process the digital signalwith fast time processing steps to generate a range spectrum comprisingN segments which correspond, respectively, to the N radio frequencyencoded transmit signals transmitted over the N transmit antennas. Inselected embodiments, the radar control processing unit is alsoconfigured to construct a MIMO virtual array by extracting informationcorresponding to the N radio frequency encoded transmit signals from theN consecutive segments in the range spectrum.

In another form, there is provided a radar system architecture andmethod for operating same. In the disclosed methodology, a transmitreference signal is generated at a transmitter module, such as bygenerating a chirp signal. In addition, the transmitter module generatesa plurality of transmit signals from the transmit reference signal, eachhaving a different frequency offset from the transmit reference signal.In selected embodiments, the plurality of transmit signals are generatedby first generating a plurality of different frequency offset tones, andthen mixing the transmit reference signal with each of the plurality offrequency offset tones to generate the plurality of transmit signals. Insuch embodiments, a frequency offset generator generates the pluralityof different frequency offset tones, and a plurality of frequency offsetmodulation mixers are connected to mix the transmit reference signalwith one of the plurality of different frequency offset tones. In otherembodiments, the plurality of transmit signals are generated byphase-shifting the transmit reference signal with a plurality offast-time phase shifters in response to a plurality of phase shiftcontrol signals to generate the plurality of transmit signals. In suchembodiments, the transmit reference signal is phase-shifted by applyingthe transmit reference signal to a bank of K-bit phase shifters tophase-shift the transmit reference signal by introducing a regularprogressive phase shift to the transmit reference signal at fast-timesampling intervals. In addition, the transmitter module encodes theplurality of transmit signals using a signal conditioning and poweramplification to produce N radio frequency encoded transmit signals. Inaddition, the transmitter module transmits the N radio frequency encodedtransmit signals over N transmit antennas. In selected embodiments, theplurality of transmit signals are encoded using time divisionmultiplexing to produce a first set of N radio frequency encodedtransmit signals for transmission over a first set of N transmitantennas during a first time slot, and to produce a second set of Nradio frequency encoded transmit signals for transmission over a secondset of N transmit antennas during a second time slot. In suchembodiments, the time division multiplexing includes activatingeven-numbered transmit antennas for transmitting the first set of Nradio frequency encoded transmit signals during the first time slot, andactivating odd-numbered transmit antennas for transmitting the secondset of N radio frequency encoded transmit signals during the second timeslot. At a first receive antenna, the receiver module receives a targetreturn signal reflected from the N radio frequency encoded transmitsignals by a target. In addition, the receiver module mixes the targetreturn signal with the transmit reference signal to produce anintermediate frequency signal. In addition, the receiver module convertsthe intermediate frequency signal to a digital signal with a high-speedanalog-to-digital converter so that the digital signal may be processedwith fast time processing steps to generate a range spectrum comprisingN segments which correspond, respectively, to the N radio frequencyencoded transmit signals transmitted over the N transmit antennas. Thedisclosed methodology may also construct a MIMO virtual array byextracting information corresponding to the N radio frequency encodedtransmit signals from the N consecutive segments in the range spectrum.

In yet another form, there is provided a computer program product storedin non-transitory machine-readable storage medium comprisinginstructions for execution by one or more processors in a radar systemhaving N transmit antennas and a receive antenna for detecting anobject. As disclosed, the computer program product includes instructionsfor configuring a reference signal generator to produce a transmitreference signal. In addition, the computer program product includesinstructions for configuring a waveform generator to produce a pluralityof transmit signals, each having a different frequency offset from thetransmit reference signal, where the plurality of transmit signals areencoded using signal conditioning and power amplification to produce andtransmit N radio frequency encoded transmit signals over N transmitantennas. In addition, the computer program product includesinstructions for configuring a downconverter to produce an intermediatefrequency signal by mixing the transmit reference signal with a targetreturn signal which is received at the receive antenna as a result ofthe N radio frequency encoded transmit signals reflecting off theobject. In addition, the computer program product includes instructionsfor configuring a high-speed analog-to-digital converter to convert theintermediate frequency signal to a digital signal. In addition, thecomputer program product includes instructions for configuring a radarcontrol processing unit to process the digital signal with fast timeprocessing steps to generate a range spectrum comprising N segmentswhich correspond, respectively, to the N radio frequency encodedtransmit signals transmitted over the N transmit antennas. In selectedembodiments, the instructions for configuring the waveform generatorinclude instructions for configuring a frequency offset signal generatorto generate a plurality of different frequency offset tones, andinstructions for configuring each of a plurality of frequency offsetmodulation mixers to mix the transmit reference signal with one of theplurality of different frequency offset tones, thereby generating afrequency offset transmit reference signal as one of the N radiofrequency encoded transmit signals. In other embodiments, theinstructions for configuring the waveform generator include instructionsfor configuring each of a plurality of fast-time phase shifters toreceive and phase-shift the transmit reference signal, therebygenerating a frequency offset transmit reference signal as one of the Nradio frequency encoded transmit signals. The computer program productmay also instructions that, when executed by the by one or moreprocessors, causes the radar system to construct a MIMO virtual array byextracting information corresponding to the N radio frequency encodedtransmit signals from the N consecutive segments in the range spectrum.The computer program product may also include instructions forconfiguring a signal encoder to encode the plurality of transmit signalsusing time division multiplexing to produce a first set of N radiofrequency encoded transmit signals for transmission over a first set ofN transmit antennas during a first time slot, and to produce a secondset of N radio frequency encoded transmit signals for transmission overa second set of N transmit antennas during a second time slot.

In still yet another form, there is provided a radar architecture,circuit, method, and system. In the disclosed system, a reference signalgenerator is configured to produce a transmit reference signal asequence of waveforms (e.g., a chirp signal). In addition, a referencesignal generator is provided and configured to produce a transmitreference signal comprising a sequence of waveforms. In addition, awaveform generator is provided and configured to produce a plurality oftransmit signals, each having a different frequency offset that is aninteger multiple of an offset frequency Δf from the transmit referencesignal. In addition, a signal encoder is provided to encode theplurality of transmit signals using a signal conditioning and poweramplification to produce N radio frequency encoded transmit signals fortransmission over N transmit antennas, where the signal encoder isconfigured to encode the plurality of transmit signals using timedivision multiplexing to produce a first set of N radio frequencyencoded transmit signals for transmission over a first set of N transmitantennas during a first time slot, and to produce a second set of Nradio frequency encoded transmit signals for transmission over a secondset of N transmit antennas during a second time slot. The system alsoincludes a receiver with at least a first receive antenna for receivinga target return signal reflected from the first and second sets of Nradio frequency encoded transmit signals by a target. The systemreceiver also includes a downconverter configured to mix the targetreturn signal with the transmit reference signal, thereby producing anintermediate frequency signal. The system receiver also includes ahigh-speed analog-to-digital converter connected to convert theintermediate frequency signal to a digital signal. In selectedembodiments, the analog-to-digital converter has a sampling rate f_(s)of at least 2N times the offset frequency Δf. Finally, the systemincludes a radar control processing unit configured to process thedigital signal with fast time processing steps to generate a rangespectrum comprising N segments which correspond, respectively, to the Nradio frequency encoded transmit signals transmitted in the first andsecond sets of N radio frequency encoded transmit signals.

Although the described exemplary embodiments disclosed herein focus onexample automotive radar circuits, systems, and methods for using same,the present invention is not necessarily limited to the exampleembodiments illustrate herein. For example, various embodiments of aco-located or distributed aperture radar may be applied innon-automotive applications, and may use additional or fewer circuitcomponents than those specifically set forth. Thus, the particularembodiments disclosed above are illustrative only and should not betaken as limitations upon the present invention, as the invention may bemodified and practiced in different but equivalent manners apparent tothose skilled in the art having the benefit of the teachings herein.Accordingly, the foregoing description is not intended to limit theinvention to the particular form set forth, but on the contrary, isintended to cover such alternatives, modifications and equivalents asmay be included within the spirit and scope of the invention as definedby the appended claims so that those skilled in the art shouldunderstand that they can make various changes, substitutions andalterations without departing from the spirit and scope of the inventionin its broadest form.

Benefits, other advantages, and solutions to problems have beendescribed above with regard to specific embodiments. However, thebenefits, advantages, solutions to problems, and any element(s) that maycause any benefit, advantage, or solution to occur or become morepronounced are not to be construed as a critical, required, or essentialfeature or element of any or all the claims. As used herein, the terms“comprises,” “comprising,” or any other variation thereof, are intendedto cover a non-exclusive inclusion, such that a process, method,article, or apparatus that comprises a list of elements does not includeonly those elements but may include other elements not expressly listedor inherent to such process, method, article, or apparatus.

What is claimed is:
 1. A radar system comprising: a reference signalgenerator configured to produce a transmit reference signal comprising asequence of waveforms; a waveform generator comprising a plurality offast-time phase shifters which are each connected to receive thetransmit reference signal and which are configured to produce aplurality of transmit signals, each having a different frequency offsetfrom the transmit reference signal; a signal encoder to encode theplurality of transmit signals using a signal conditioning and poweramplification to produce and transmit N radio frequency encoded transmitsignals over N transmit antennas; at least a first receive antenna forreceiving a target return signal reflected from the N radio frequencyencoded transmit signals by a target; a downconverter configured to mixthe target return signal with the transmit reference signal, therebyproducing an intermediate frequency signal; an analog-to-digitalconverter connected to convert the intermediate frequency signal to adigital signal; and a radar control processing unit configured toprocess the digital signal with fast time processing steps to generate arange spectrum comprising N segments which correspond, respectively, tothe N radio frequency encoded transmit signals transmitted over the Ntransmit antennas.
 2. The radar system of claim 1, where the radarcontrol processing unit is further configured to construct a MIMOvirtual array by extracting information corresponding to the N radiofrequency encoded transmit signals from the N consecutive segments inthe range spectrum.
 3. The radar system of claim 1, where theanalog-to-digital converter has a sampling rate of at least N times ofthe minimally required sampling rate of a single-transmitter system thatdetects targets unambiguously in range.
 4. The radar system of claim 1,where each of the plurality of fast-time phase shifters is connected toreceive the transmit reference signal and to generate a frequency offsetmodulated encoded transmit signals.
 5. The radar system of claim 1,where the plurality of fast-time phase shifters comprises a bank ofK-bit phase shifters for introducing a regular progressive phase shiftto the transmit reference signal at fast-time sampling intervals.
 6. Theradar system of claim 1, where the signal encoder is configured toencode the plurality of transmit signals using time divisionmultiplexing to produce a first set of N/2 radio frequency encodedtransmit signals for transmission over a first set of N/2 transmitantennas during a first time slot, and to produce a second set of N/2radio frequency encoded transmit signals for transmission over a secondset of N/2 transmit antennas during a second time slot.
 7. The radarsystem of claim 6, where the signal encoder is configured to encode theplurality of transmit signals using time division multiplexing byactivating even-numbered transmit antennas for transmitting the firstset of N/2 radio frequency encoded transmit signals during the firsttime slot, and by activating odd-numbered transmit antennas fortransmitting the second set of N/2 radio frequency encoded transmitsignals during the second time slot.
 8. The radar system of claim 1,further comprising a notch filter bank connected between thedownconverter and the analog-to-digital converter for filtering theintermediate frequency signal to remove zero-range interference in afast-time spectrum of each transmitter.
 9. The radar system of claim 1,where the downconverter comprises an I/Q mixer which is connected tofirst and second analog-to-digital converters which respectively convertthe intermediate frequency signal to an I-channel digital signal andQ-channel digital signal.
 10. A method for operating a radar system,comprising: generating a transmit reference signal comprising a sequenceof chirp waveforms at a transmitter module; generating a plurality oftransmit signals from the transmit reference signal at the transmittermodule using a plurality of fast-time phase shifters which are connectedto receive the transmit reference signal and configured to produce aplurality of transmit signals, each having a different frequency offsetfrom the transmit reference signal; encoding the plurality of transmitsignals using a signal conditioning and power amplification at thetransmitter module to produce N radio frequency encoded transmitsignals; transmitting the N radio frequency encoded transmit signalsover N transmit antennas at the transmitter module; receiving, at leasta first receive antenna of a receiver module, a target return signalreflected from the N radio frequency encoded transmit signals by atarget; mixing the target return signal with the transmit referencesignal at the receiver module to produce an intermediate frequencysignal; converting the intermediate frequency signal to a digital signalwith an analog-to-digital converter at the receiver module; andprocessing the digital signal with fast time processing steps togenerate a range spectrum comprising N segments which correspond,respectively, to the N radio frequency encoded transmit signalstransmitted over the N transmit antennas.
 11. The method of claim 10,further comprising constructing a MIMO virtual array by extractinginformation corresponding to the N radio frequency encoded transmitsignals from the N consecutive segments in the range spectrum.
 12. Themethod of claim 10, where generating the plurality of transmit signalscomprises: generating a plurality of different frequency offset tones;and mixing the transmit reference signal with each of the plurality offrequency offset tones to generate the plurality of transmit signals.13. The method of claim 12, where a frequency offset generator generatesthe plurality of different frequency offset tones, and where a pluralityof frequency offset modulation mixers are connected to mix the transmitreference signal with one of the plurality of different frequency offsettones.
 14. The method of claim 10, where generating the plurality oftransmit signals comprises phase-shifting the transmit reference signalwith the plurality of fast-time phase shifters in response to aplurality of phase shift control signals to generate the plurality oftransmit signals.
 15. The method of claim 14, where phase-shifting thetransmit reference signal comprises applying the transmit referencesignal to a bank of K-bit phase shifters to phase-shift the transmitreference signal by introducing a regular progressive phase shift to thetransmit reference signal at fast-time sampling intervals.
 16. Themethod of claim 10, where encoding the plurality of transmit signalscomprises encoding the plurality of transmit signals using time divisionmultiplexing to produce a first set of N/2 radio frequency encodedtransmit signals for transmission over a first set of N/2 transmitantennas during a first time slot, and to produce a second set of N/2radio frequency encoded transmit signals for transmission over a secondset of N/2 transmit antennas during a second time slot.
 17. The methodof claim 16, where encoding the plurality of transmit signals using timedivision multiplexing comprises activating even-numbered transmitantennas for transmitting the first set of N/2 radio frequency encodedtransmit signals during the first time slot, and activating odd-numberedtransmit antennas for transmitting the second set of N/2 radio frequencyencoded transmit signals during the second time slot.
 18. A radar systemcomprising: a reference signal generator configured to produce atransmit reference signal comprising a sequence of chirp waveforms; awaveform generator comprising a plurality of fast-time phase shifterswhich are each connected to receive the transmit reference signal andwhich are configured to produce a plurality of transmit signals, eachhaving a different frequency offset that is an integer multiple of anoffset frequency Af from the transmit reference signal; a signal encoderto encode the plurality of transmit signals using a signal conditioningand power amplification to produce N radio frequency encoded transmitsignals for transmission over N transmit antennas, where the signalencoder is configured to encode the plurality of transmit signals usingtime division multiplexing to produce a first set of N radio frequencyencoded transmit signals for transmission over a first set of N transmitantennas during a first time slot, and to produce a second set of Nradio frequency encoded transmit signals for transmission over a secondset of N transmit antennas during a second time slot; at least a firstreceive antenna for receiving a target return signal reflected from thefirst and second sets of N radio frequency encoded transmit signals by atarget; a downconverter configured to mix the target return signal withthe transmit reference signal, thereby producing an intermediatefrequency signal; an analog-to-digital converter connected to convertthe intermediate frequency signal to a digital signal; and a radarcontrol processing unit configured to process the digital signal withfast time processing steps to generate a range spectrum comprising Nsegments which correspond, respectively, to the N radio frequencyencoded transmit signals transmitted in the first and second sets of Nradio frequency encoded transmit signals.
 19. The radar system of claim18, where the analog-to-digital converter has a sampling rate f_(s) ofat least 2N times the offset frequency Δf.